History Of The Transistor.doc 3253d

History of the Transistor by Tony van Roon (VA3AVR)

Dr. John Bardeen(left), Dr. Walter Brattain(right), and Dr. William Shockley(center) discovered the transistor effect and developed the first device in December, 1947, while the three were of the technical staff at Bell Laboratories in Murray Hill, NJ. They were awarded the Nobel Prize in physics in 1956. The PN Junction: What the Bell Labs scientists discovered was that silicon was comprised of two distinct regions differentiated by the way in which they favored current flow. The area that favored positive current flow they named "P" and the area that favored negative current flow they named "N." More importantly, they determined the impurities that caused these tendencies in the "P" and "N" regions and could reproduce them at will. With the discovery of the P-N junction and the ability to control its properties, the fundamental ground work was laid for the invention of the transistor. This Bell Labs discovery was instrumental in the development of all semiconductor devices to come. The First Silicon Transistor: It was late afternoon in 1954 at a conference for the Institute of Radio Engineers. Many people giving talks had complained about the current germanium transistors--they had a bad habit of not working at high temperatures. Silicon, since it's right above germanium on the periodic table and has similar properties, might make a better gadget. But, they said, no one should expect a silicon transistor for years. Then Gordon Teal of Texas Instruments stood up to give his talk. He pulled three small objects out of his pocket and announced: "Contrary to what my colleagues have told you about the bleak prospects for silicon transistors, I happen to have a few of them here in my pocket." That moment catapulted TI from a small start-up electronics company into a major player. They were the first company to produce silicon transistors -- and consequently the first company to produce a truly consistent mass-produced transistor. Scientists knew about the problems with germanium transistors. Germanium worked, but it had its mood swings. When the germanium heated up--a natural outcome of being part of an electrical circuit--the transistor would have too many free electrons. Since a transistor only works because it has a specific, limited amount of electrons running around, high heat could stop a transistor from working altogether. While still working at Bell Labs in 1950, Teal began growing silicon crystals to see if

they might work better. But just as it had taken years to produce pure enough germanium, it took several years to produce pure enough silicon. By the time he succeeded, Teal was working at Texas Instruments. Luring someone as knowledgeable about crystals as Teal away from Bell proved to be one of the most important things TI ever did. On April 14, 1954, Gordon Teal showed TI's Vice President, Pat Haggerty, a working silicon transistor. Haggerty knew if they could be the first to sell these new transistors, they'd have it made. The company jumped into action -- four weeks later when Teal told his colleagues about the silicon transistors in his pocket, TI had already started production. The Invention of the First Transistor November 17 - December 23, 1947 Getting 'Wet': On November 17, 1947, Walter Brattain dumped his whole experiment into a thermos of water. The silicon contraption he'd built was supposed to help him study how electrons acted on the surface of a semiconductor -- and why whatever they were doing made it impossible to build an amplifier. But condensation kept forming on the silicon and mes the experiment. To get rid of that condensation, Brattain probably should have put the silicon in a vacuum, but he decided that would take too long. Instead he just dumped the whole experiment under water -- it certainly got rid of the condensation! Out of the blue, the wet device created the largest amplification he'd seen so far. He and another scientist, Robert Gibney, stared at the experiment, stunned. They began fiddling with different knobs and buttons: by turning on a positive voltage they increased the effect even more; turning it to negative could get rid of it completely. It seemed that whatever those electrons had been doing on the surface to block amplification had somehow been canceled out by the water--the greatest obstacle to building an amplifier had been overcome. Putting the Idea to Use. When John Bardeen was told what had happened he thought of a new way to make an amplifier. On November 21, Bardeen suggested pushing a metal point into the silicon surrounded by distilled water. The water would eliminate that exasperating electron problem just under the point as it had in the thermos. The tough part was that the point couldn't touch the water, it must only touch the silicon. But as always, Brattain was a genius in the lab. He could build anything. And when this amplifier was built, it worked. Of course, there was only a tiny bit of amplification--but it worked. Big Amplification.

Once they'd gotten slight amplification with that tiny drop of water, Bardeen and Brattain figured they were on the road to something worthwhile. Using different materials and different setups and different electrolytes in place of the water, the two men tried to get an even bigger increase in current. Then on December 8, Bardeen suggested they replace the silicon with germanium. They got a current jump, all right--an amplification of some 330 times--but in the exact opposite direction they'd expected. Instead of moving the electrons along, the electrolyte was getting the holes moving. But amplification is amplification -it was a start. Brattain Makes a Mistake Unfortunately this giant jump in amplification only worked for certain types of current -ones with very low frequencies. That wouldn't work for a phone line, which has to handle all the complex frequencies of a person's voice. So the next step was to get it to work at all kinds of frequencies. Bardeen and Brattain thought it might be the liquid which was the problem. So they replaced it with germanium dioxide -- which is essentially a little bit of germanium rust. Gibney prepared a special slab of germanium with a shimmering green oxide layer on one side. On December 12, Brattain began to insert the point s. Nothing happened: In fact the device worked as if there was no oxide layer at all. And as Brattain poked the gold in again and again, he realized that's because there wasn't an oxide layer. He had washed it off by accident. Brattain was furious with himself, but decided to fiddle with the point anyway. To his surprise, he actually got some voltage amplification -- and more importantly he could get it at all frequencies! The gold was putting holes into the germanium and these holes canceled out the effect of the electrons at the surface, the same way the water had. But this was much better than the version that used water, because now, the device was increasing the current at all frequencies. Bringing it All Together: In the past month, Bardeen and Brattain had managed to get a large amplification at some frequencies and they'd gotten a small amplification for all frequencies -- now they just had to combine the two. They knew that the key components were a slab of germanium and two gold point s just fractions of a millimeter apart. Walter Brattain put a ribbon of gold foil around a plastic triangle, and sliced it through at one of the points. By putting the point of the triangle gently down on the germanium, they saw a fantastic effect -- signal came in through one gold and increased as as it raced out the other. The first point- transistor had been made. Telling the Brass: For a week, the scientists kept their success a secret. Shockley asked Bardeen and Brattain to show off their little plastic triangle at a group meeting to the lab and the higher-ups on December 23. After the rest of the lab had a chance to look it over and

conduct a few tests, it was official -- this tiny bit of germanium, plastic and gold was the first working solid state amplifier. --------------Shockley Invents the Junction Transistor January and February, 1948 A Solitary New Year's Eve William Shockley spent New Year's Eve alone in a hotel in Chicago. He was there for a Physical Society meeting, but he was most excited about having some time to himself to concentrate on his work. There may have been a party going on downstairs, but Shockley wanted nothing to do with it. He had more important things to think about. He spent that night and the next two days working on some of his ideas for a new transistor-one that would improve on Bardeen and Brattain's ideas. Scratching page after page into his notebook, one of Shockley's ideas was to build a semiconductor "sandwich." Three layers of semiconductors all piled together, he thought, just might work like a vacuum tube-with the middle layer turning current on and off at will. After some 30 pages of notes, the concept hadn't quite come together so Shockley set it aside to do other work. The Idea Comes Together Shockley's January was pretty dismal. He thought he should get sole credit for inventing the transistor--the initial research ideas, after all, had been his own. The Bell Labs attorneys didn't agree. They refused to even put him on the patent. The only thing to do, Shockley decided, was to build a better mouse trap. As the rest of the group worked merrily away on improving Brattain and Bardeen's pointjunction transistor. Shockley concentrated on his own ideas -- never letting anyone else in the lab know what he was up to. On January 23, unable to sleep, Shockley was sitting at the kitchen table bright and early in the morning. He suddenly had a revelation. Building on the "sandwich" device he'd come up with on New Year's Eve, he thought he had an idea for an improved transistor. This would be three-layered sandwich. The outermost pieces would be semiconductors with too many electrons, while the bit in the middle would have too few electrons. The middle layer would act like a faucet--as the voltage on that part was adjusted up and down, it could turn current in the sandwich on and off at will. Shockley told no one about his idea. The physics behind this amplifier was very different from Bardeen's and Brattain's, since it involved current flowing directly through the chunks of semiconductors, not along the surface. No one was sure if current even could flow right through a semiconductor and possibly Shockley wanted to test it before discussing it. Or possibly he felt that Bardeen and Brattain had "taken" ideas of his for the

point- transistor and he didn't want to risk that happening again. The Eureka Moment Then, on February 18, Shockley learned it could work. Two of the group, Joseph Becker and John Shive, were working on a separate experiment. Their results could only be explained if the electrons did in fact travel right through the bulk of a semiconductor. When they presented their findings to the group, Shockley knew he had the proof he needed. He jumped up and for the first time shared his concept of a sandwich transistor to the rest of his team. Bardeen and Brattain were stunned that they hadn't been filled in before now. It was clear that Shockley had been keeping this secret for weeks. It added still more space to the ever-widening gap that was growing between them. --------------Telling the Military June 23, 1949 They had no way of knowing all that the transistor could do, but the s at Bell Labs still knew they were on to something big. They were about to hold a huge press conference to announce what they'd invented -- but before telling the public they had to check with the military. At the very least, the transistor could revolutionize communications and radio signals, something that would give the US Army an advantage if the invention was kept a secret from other countries. Bell's president, Mervin Kelly, hoped the army wouldn't want to classify this research, but he knew it just might happen. On June 23, Ralph Bown gave a presentation to a group of military officers. He showed the way the tiny bit of crystal and wire could amplify an electrical signal much more efficiently than a bulky vacuum tube could. He also told them this was the same demonstration he was preparing to give to the press the next week. What he didn't do was ask permission. Bown and Kelly didn't want to make it easy for the military to classify the transistor. If they wanted to keep it a secret, the army would have to bring up the subject itself. The armed services went home to their various offices and discussed whether to classify Bell's work. There were certainly those who thought that, at the very least, it should be kept secret until it was better understood just what the transistor could do. But in the end, nobody said a word. Bell Labs went on to its big press conference without a hitch. --------------A Working Junction Transistor 1948-1951 There was no doubt about it, point- transistors were fidgety. The transistors being made by Bell just didn't work the same way twice, and on top of that, they were noisy.

While one lab at Bell was trying to improve those first type-A transistors, William Shockley was working on a whole different design that would eventually get rid of these problems. Early in 1948, Shockley conceived of a transistor that looked like a sandwich, with two layers of one type of semiconductor surrounding a second kind. This was a completely different setup which didn't have the shaky wires that made the point- transistors so hard to control. Not Just on the Surface A working sandwich transistor would require that electricity travel straight across a crystal instead of around the surface. But Bardeen's theory about how the point- transistor worked said that electricity could only travel around the outside of a semiconductor crystal. In February of 1948, some tentative results in the Shockley lab suggested this might not be true. So the first thing Shockley had to do was determine just what was going on. Careful experiments led by a physicist in the group, Richard Haynes, helped. Haynes put electrodes on both sides of a thin germanium crystal and took very sensitive measurements of the size and speed of the current. Electricity definitely flowed straight through the crystal. That meant Shockley's vision of a new kind of transistor was theoretically possible. Growing Crystals But Haynes also discovered that the layer in the middle of the sandwich had to be very thin and very pure. The man who paved the way for growing the best crystals was Gordon Teal. He didn't work in Shockley's group, but he kept tabs on what was going on. He'd even been asked to provide crystals for the Solid State team upon occasion. Teal thought transistors should be built from a single crystal-as opposed to cutting a sliver from a larger ingot of many crystals. The boundaries between all the little crystals caused ruts that scattered the current, and Teal had heard of a way to build a large single crystal which wouldn't have all those crags. The method was to take a tiny seed crystal and dip it into the melted germanium. This was then pulled out ever so slowly, as a crystal formed like an icicle below the seed. Teal knew how to do it, but no one was interested. A number of institutions at the time, Bell included, had a bad habit of not trusting techniques that hadn't been devised at home. Shockley didn't think these single crystals were necessary at all. Jack Morton, head of the transistor-production group, said Teal should go ahead with the research, but didn't throw much his way. Luckily, Teal did continue the research, working with engineer John Little. Three months

later, in March of 1949, Shockley had to it he'd been wrong. Current flowing across Teal's semiconductors could last up to one hundred times longer than it had in the old cut crystals. Growing Even Better Crystals Nice crystals are all well and good, but a sandwich transistor needed a sandwich crystal. The outer layers had to be a semiconductor with either too many electrons (known as Ntype) or too few (known as P-type), while the inner layer was the opposite. Under Shockley's prodding, Teal and Morgan Sparks began adding impurities to the melt while they pulled the crystal out of the melt. Adding impurities is known as "doping," and it's how one turns a semiconductor into N- or P-type. As they pulled the seed crystal out of an N-type germanium melt, they quickly added some gallium to turn the melt into P-type. As a layer of P-type formed on the everlengthening crystal, they added antimony, which compensated for the gallium and turned the melt back into N-type. Once the process was done, there was a single, thin crystal formed into a perfect sandwich. By etching away the surface of the outside layers, Sparks and Teal left a tiny bit of P-type crystal protruding. To this they attached a fine electrode-creating a circuit the way Shockley had envisioned. On April 12, 1950, they tested what they had built. Without a doubt, more current came out of the sandwich than went in. It was a working amplifier. The First Junction Transistor The first junction transistor had been born. But It Wasn't a Very Good One . . . Yet This transistor could amplify electrical signals, but not particularly complicated ones. If the signal changed rapidly, as a voice coming over a phone line does, the transistor couldn't keep up and would garble the output. The problem lay in the middle of the sandwich: it was too easy for electric current to spread out and become unfocused as it crossed the P-type layer. To solve the problem, the layer had to be even thinner. In January of 1951, Morgan Sparks figured out a way to accomplish that. By pulling the crystal out more slowly than ever, while constantly stirring the melt, he managed to get the middle layer of the sandwich thinner than a sheet of paper. This new, improved sandwich did all that the researchers hoped. They still weren't up to the point- transistor's ability to handle signals that fluctuated extremely rapidly, but in every other way they were superior. They were much more efficient, used very little power to work, and they were so much quieter that they could handle weaker signals than the type-A transistors ever could.

In July of 1951, Bell held another press conference -- this time announcing the invention of a working and efficient junction transistor. --------------Sharing the Technology: Bell Hosts Transistor Symposia 1951-1952 Bell Labs had an important realization: development of the transistor was going to move a lot more quickly if they opened up the field to other companies. So in September 1951, Bell Labs hosted a symposium to spread the gospel about what the transistor could do. Attending the conference were some 300 scientists and engineers. The attendees all went home to their respective companies with a great sense of what the transistor could do -but little idea of how to build one. For that knowledge, Bell announced, a company would have to pay a licensing fee of $25,000. Twenty-six companies, from both the US and abroad, signed up for the privilege. The companies were both big, such as IBM and General Electric, and small, such as then-unknowns like Texas Instruments. Over one hundred registrants from the select companies returned for the Transistor Technology Symposium in April of 1952. For eight days Bell Labs worked the attendees day and night -- but at the end, they were equipped to go off and build transistors for themselves. Bell took all the information from the meeting and bound it into a two volume book set called "Transistor Technology." The book became fondly known as "Mother Bell's Cookbook." ---------------William Shockley Moves to California 1956 William Shockley had gone as far as he was going to go at Bell Labs. He had watched the people underneath him get promoted above him -- and with good reason. Too many top quality scientists hadn't been able to work with him . A genius he may have been, but a good manager he was not. Shockley decided he needed a big change. The first thing to go was the car -- he traded in the fancy MG for a Jaguar convertible. Next: the job. He spent a semester at Caltech and then a year working for the Weapons Systems Evaluation Group in Washington DC., but nothing completely satisfied him. Eager to be able to run things his own way, he finally decided to strike out on his own -- get some funding and start his own company. In August of 1955, Shockley flew to LA to spend a week with his new friend Arnold Beckman, a California chemist and businessman. Shockley shared his dream of starting a company to build cutting edge semiconductor devices. Beckman was sold on the idea and agreed to underwrite the venture.

Shockley was lured to the Palo Alto area by Stanford's provost, Fred Terman who thought that a solid research institution in the area would benefit Stanford. With a location picked out, Shockley just had to find the people. He wanted to staff his company with only the best and the brightest. He first sought to employ his colleagues from Bell Labs, but they wouldn't make the jump to the west coast -- or perhaps they couldn't make the jump to working with Shockley again. So Shockley began traveling all over the country recruiting young scientists. At a lavish luncheon in February of 1956, Shockley and Beckman announced the formation of their brand new lab. They only had four employees at the time, but Shockley Semiconductor Laboratory had officially opened for business. Shockley's was the first company of its kind to settle in the Palo Alto area, but over the years more and more semiconductor labs -- and the computer industries they initiated -- flocked to the area. It wasn't long before the region had earned a new name: Silicon Valley. --------------The Future of Transistors The first announcement of the invention of the transistor met with almost no fanfare. The integrated circuit was originally thought to be useful only in military applications. The microprocessor's investors pulled out before it was built, thinking it was a waste of money. The transistor and its offspring have consistently been under valued -- yet turned out to do more than anyone predicted. Today's predictions also say that there is a limit to just how much the transistor can do. This time around, the predictions are that transistors can't get substantially smaller than they currently are. Then again, in 1961, scientists predicted that no transistor on a chip could ever be smaller than 10 millionth of a meter -- and on a modern Intel Pentium chip they are 100 times smaller than that. With hindsight, such predictions seem ridiculous, and it's easy to think that current predictions will sound just as silly thirty years from now. But modern predictions of the size limit are based on some very fundamental physics -- the size of the atom and the electron. Since transistors run on electric current, they must always, no matter what, be at least big enough to allow electrons through. On the other hand, all that's really needed is a single electron at a time. A transistor small enough to operate with only one electron would be phenomenally small, yet it is theoretically possible. The transistors of the future could make modern chips seem as big and bulky as vacuum tubes seem to us today. The problem is that once devices become that tiny, everything moves according to the laws of quantum mechanics -- and quantum mechanics allows electrons to do some weird things. In a transistor that small, the electron would act more like a wave than a single particle. As a wave it would smear out in space, and could even tunnel its way through the transistor without truly acting on it. Researchers are nevertheless currently working on innovative ways to build such tiny

devices -- abandoning silicon, abandoning all of today's manufacturing methods. Such transistors are known, not surprisingly, as single electron transistors, and they'd be considered "on" or "off" depending on whether they were holding an electron. (Transistors at this level would be solely used as switches for binary coding, not as amplifiers.) In fact, such a tiny device might make use of the quantum weirdness of the ultra-small. The electron could be coded to have three positions -- instead of simply "on" or "off" it could also have "somewhere between on and off. This would open up doors for entirely new kinds of computers. At the moment, however, there are no effective single electron transistors. Even without new technologies, there's room for miniaturization. By improving on current building techniques, it's likely that current transistors will be at least twice as small by 2010. With nearly a billion transistors on Intel's latest processor that would mean four times as many transistors on a chip are theoretically possible. Chips like this would allow computers to be much "smarter" than they currently are. ---------------#--------------That was quite a bit of interesting history of the transistor! Let's move on to the tid bids of the Bipolar Junction Transistor (BJT for short), mostly the common 2N3904 type in this exciting 9-part tutorial. Copyright and Credits Tutorial, photos, drawings, schematics, etc., copyright © 2006, Tony van Roon, unless otherwise indicated. The content of this article is a collection of excerpts found on the web and re-posted with permission. Page copyright © 2006 - Tony van Roon, VA3AVR Last updated: January 13, 2007

Transistors Tutorial Part 1: "Bipolar Basics"

"We look at the tiny devices that have reshaped the world of electronics." Along with the solid-state diode, the point- transistor--invented in 1947 at Bell Labs--started the semiconductor revolution and has gone on the become one of the rudimentary devices in today's electronic equipment. The transistor, whether in discrete or IC form, is at the heart of most modern circuitry. Therefore, understanding how transistors function will help you properly design circuits containing them, and in case of a failure, enable you to find and correct the problem. Bipolar-Transistor Composition: A bipolar transistor is basically a two PN junctions connected back-to-back within the same piece of semiconductor material and sharing a common P- or N-doped semiconductor region. There are two types of bipolar transistor, the NPN and the PNP. Fig. 1A is a simplified illustration of the composition of the NPN type of transistor. In our illustration, the NPN type unit is shown as P-doped semiconductor material sandwiched between two layers of N-doped material. The composition of a PNP transistor is just the opposite of that, (i.e. the N- and P-doped materials in the transistor are interchanged). It follows then that biasing considerations for NPN units are also opposite from those for the PNP unit. Note from Fig. 1A that a bipolar transistor is comprised of a center region called the base surrounded by two other regions known as the collector and the emitter. The difference between them will be discussed shortly. The two junctions are arranged so that they are very close together; that's done by making the shared base region very thin and lightly doped. That causes the two junctions to interact with one another. Conduction is the collector-base junction depends largely on what happens in the emitter-base junction. Because the region is lightly doped, it has a relatively small number of free carriers (holes in a P-type base and electronics in an N-type base) to conduct current. On the other hand, the emitter region is quite heavily doped, containing a much larger amount of donor impurity (for the NPN type) or acceptor impurity (for the PNP type), so there are many more free carriers available in the emitter region to conduct current than in the adjacent base region. Because of that, the emitter-base junction, when forward biased, conducts

much the same as a common PN junction diode. The current that flows (composed of electrons for NPN units and holes, in the case of PNP transistors) is mainly from the emitter to the base rather than vice versa. That is where the emitter derives its name--it emits or injects current carriers in the other regions of the device. The third region of a transistor, the collector, is lightly doped, much the same as the base, except with the opposite type of doping impurity, so it (like the base region) has relatively few free carriers available to conduct current in the normal way. The collector-base junction is normally reverse biased, so a depletion layer forms, spreading out on either side of the junction. The depletion layer effectively removes the carriers that would otherwise balance out the charges on the fixed impurity atoms of the crystals, setting up a potential barrier to match the applied reverse voltage.

To the normal majority carriers in the base and emitter, that potential barrier is a big wall that must be overcome before they can to the other side. So just as in the case of a normal diode, virtually no current flows across the collector-base junction when left to its own devices. However, the junction is not left to its own devices. that the base region is deliberately made very thin and lightly doped, while

the emitter is made much more heavily doped. Because of that, applying a forward bias to the emitter-base junction causes vast majority carriers to be injected into th the base, and straight into the reverse-biased collector-base junction. Those carriers are actually minority carriers in the base region, because that region is of opposite semiconductor type to the emitter. To those majority-turned-minority carriers, the collector-base junction depletion region is not a barrier at all but an inviting, accelerating filed; so as soon as they reach the depletion layer, they are immediately swept into the collector region. Forward biasing the emitter-base junction causes two things to happen that might seem surprising at first: Only a relatively small current actually flows between the emitter and the base. much smaller than would flow in a normal PN diode despite the forward bias applied to the junction between them. A much larger current instead flows directly between the emitter and the collector regions, in this case, despite the fact that the collector-base junction is reversed biased. That effect is illustrated in Fig. 1A, which (hopefully) will help you to understand what is going on. The diagram shows a NPN transistor, but the action in a PNP unit is similar except for the opposite region polarity and conduction mainly by holes rather than electrons. From a practical point of view, the behavior of bipolar transistors means that, unlike the simple PN-junction diode, it is capable of amplification. In effect, a small input current made to flow between the emitter and collector. Only a small voltage--around 0.6 volts for a typical silicon transistor--is needed to produce the small input current required. In contrast, the reverse-bias voltage applied across the collector-base junction can be much larger; typically anywhere from 6 to 90 volts or more. So in producing and being able to control a larger current in this much higher-output circuit, the transistor's small input current and voltage can achieve considerable voltage, power, and current, gains. Bipolar transistors, therefore, work very well as both amplifiers and electronics switches. That is why they have become the workhorses of modern electronics, virtually replacing the vacuum tube. The diagram in Fig. 1A is designed to show how a bipolar transistor works, rather than its physical construction. The actual form of the modern, planar, double-defuse epitaxial-junction transistor is shown in Fig. 1A. The collector region is formed from a lightly doped layer grown epitaxially on the main substrate, which is made from the same type (but more heavily doped) material to provide a low resistance connection. Here, both are N-doped material; for a PNP transistor, they would be P-doped material. The base region is formed by lightly diffusing the opposite type impurity into a mediumsized area of the chip surface to reverse that type of area and create the base-collector unction. The emitter region is formed by a second and heavier diffusion over the smaller area inside the first, but this time with the same kind of impurity as used for the epitaxial collector region. The second diffusion is very carefully controlled so that the emitter region that results extends almost--but not quite--to the bottom of the base. That leaves the area of the base right below the emitter quite thin to ensure that as many as possible of the carriers are injected from the emitter region will be swept through to the collector. The thinner that active base region, the higher (in general) the gain of the transistor. Note that although the collector and emitter regions are made of the same type of semiconductor material, the two are physically quite different. The emitter is heavily

doped (for a good carrier injection) and can be relatively small since the emitter-base junction does not need to dissipate much power (heat). In contrast, the collector is lightly doped (for a wide depletion area) and its junction is much larger since, being reversed biased, it must dissipate much more power. Connections to the emitter and base regions are made by way of aluminum electrodes deposited on the surface. Thin wires are bonded to the electrodes for connection to the main device leads. The low-resistance substrate itself is used to connect to the collector region. That is the basic construction used for most modern bipolar transistors, whether they are discrete units or part of an IC containing thousands of transistors. The main difference is size, although, in an IC, the collector region of the transistor will generally be in an epitaxial layer grown on the opposite kind of substrate, and separated by diffused walls (of the opposite type material) to separate the transistors from each other. In an IC, the active part of an individual transistor might only be a couple micrometers square, while a very large transistor (used to switch hundreds of amperes) might be on a single wafer of 10 mm or more in diameter. Typical small-to-medium power, discrete transistors used in consumer and hobby electronics are grown on chips measuring from 1to about 3-mm square--the rest of the component is protective packaging. Transistor Operation: Refer to Fig. 2, a PNP version of the illustration shown in Fig. 1A. Note that both are essentially the same, except that in this instance, the collector is more negative than the base or the emitter. That is an important characteristic to when it comes to the operation of bipolar transistors. If a positive voltage is applied to the P-doped emitter (to the left), current will be swept through the base-emitter junction--with the holes from the P-doped material moving to the right and the electrons form the N-doped material moving to the left. Some of the holes moving into the N-doped base region will combine with the electrons and become neutralized, while others will migrate to the base-collector junction. Normally, if the base-collector junction is negatively biased, there would be no current flow in the circuit. However, there would be additional holes in the junction to travel to the base-collector junction, and electrons can then travel toward the base-emitter junction, so a current flows even through that section of the sandwich is biased (at cutoff) to prevent conduction. Most of the current travels between the emitter and collector and does not flow out through the base. The amplitude of the collector current depends principally on the magnitude of emitter current (e.g., the collector current). Note that between each PN junction, there is an area known as the depletion or transition region that is similar in some characteristics to a dielectric layer. That layer varies in accordance with the operating voltage. The semiconductor materials on either side of the depletion regions constitute the plates of a capacitor. The base-collector capacitance is indicated in Fig. 2 as Cbc, and the baseemitter capacitance is designated Cbe. A change in signal and operating voltages causes a

non-linear change in those junction capacitances. There is also a base-emitter resistance (Rbe that must be considered. In practical transistors, emitter resistance is on the order of a few ohms, while the collector resistance is many hundreds or even thousands of times larger. The junction capacitance in combination with the base-emitter resistance determine the useful upper-frequency limit of a transistor by establishing an RC time constant. Because the collector is reversed biased, the collector-to-base resistance is high. On the other hand, the emitter and collector currents are substantially equal, so the power in the collector circuit is larger than the power in the emitter circuit. (P = I2R, so the powers are proportional to the respective resistances, if the currents are the same.) In practical transistors, emitter resistance is on the order of a few ohms, while the collector resistance is many hundreds or thousands of times larger, so power gains of 20 to 40dB, or even more, are possible. Figure 3 shows the schematic symbols for both the NPN and PNP version of the bipolar transistor. The first two letters of the designators (NPN or PNP) indicate the polarities of the voltages applied to the collector and emitter in normal operation. For example, in a PNP unit, the emitter is made more positive with respect to the collector and the base, and the collector is made more negative with respect to the base. Another way of saying that is: the collector is more negative than the base and the base is more negative than the emitter. Transistor Amplifiers: Transistors are among the most commonly used building blocks in electronics. While they can be used as electronically controlled switches, they are widely configured for amplifier use. In fact, the vast majority of electronic circuits contain one or more amplifiers of some type or another. However, what exactly do we mean by the term amplifier? By definition an amplifier is a circuit that draws power from a source other than the input signal and produces an output that is usually an enlarged reproduction of the input signal. We say usually because not all amplifiers are used to magnify the input signal--buffer amplifiers (often called unity-gain amplifiers) are not designed to magnify the input signal. When operated as a buffer, the transistor is used to isolate one stage from the effects of one that follows. Since buffer amplifiers provide no increase in signal level, a 10-millivolt (mV) signal applied to the input of a unity-gain amplifier produces an output signal at the same 10-mV level (a carbon copy of the input signal). There are may types of amplifiers, however, and all fall into one of two broad categories: voltage amplifiers or current (often referred to as a power) amplifiers. The term voltage amplifier implies to a circuit in which a low voltage is applied to the input to produce a higher voltage at the output. The term power amplifier is generally reserved for those that supply an appreciable power (or current) increase to the load.

Because a vast array of amplifier circuits in use in modern electronics, amplifier circuits are often subdivided by application--AF, IF, RF, Instrumentation, op-amp, etc. Another way of categorizing amplifiers is by configuration: common-emitter, common-collector, and common-base for example. The important parameters in such circuits are the cutoff frequency and the input/output impedances. The cut-off frequency at which the gain of an amplifier falls below 0.707 times the maximum gain of the circuit. The input impedance is the output impedance of the transistor.

Amplifier Configurations: An example of a common-base amplifier is shown in Fig. 4A. The optimum load impedance can range from a few thousand ohm to 100,000 ohms, depending on the circuit's requirements. In this type of circuit, the output signal (at the collector) is in phase with the input signal (applied at the emitter). THe current that flows through the base resistance of the transistor is therefore in phase as well, so the circuit tends to be regenerative and will oscillate if the current-amplification factor is greater than one. A common-emitter (also called a "grounded-emitter") amplifier is shown in Fig. 4B. Base current in this amplifier configuration small and the input impedance is therefore fairly high (several thousand ohms on the average). Collector resistance on the other hand, can

be tens of thousands of ohms, depending on the signal's source impedance. The commonemitter amplifier has a lower cutoff frequency than does the common-base type, but gives the highest power gain of the three configurations. Note that the output signal is 180° outof-phase with (or the opposite of) the input (base-current) signal, so the that flows through the small emitter resistance is negative (degenerative), keeping the circuit stable. The common-emitter amplifier is one of the most often seen configurations for the bipolar transistor. The common-collector amplifier (also referred to as an emitter follower), see Fig. 4C, has a high input impedance and a low output impedance. The impedance is approximately: The fact that the input resistance is directly related to the load resistance is a disadvantage of this type of amplifier if the load is one whose resistance or impedance varies with frequency. The current transfer ratio of this type of circuit is: and the cutoff frequency is the same as in the common-emitter amplifier circuit. The output and input currents of this type of circuit are in phase.

Amplifier Classifications: Amplifiers may be otherwise classified by their specific operational characteristics, in particular, the bias voltages between the emitter-base and base-collector junctions. The relationship between the bias voltage and the cutoff voltage of an amplifier is what classifies an amplifier as being class A, B, C, or AB. Each class has a specific characteristic that makes it most suitable for a particular application. In a class-A amplifier--which is the least efficient, but offers the least distortion--the transistor is biased so that its quiescent operation point is in the middle of the powersupply extremes, i.e., the transistor is always turned on and the resulting output varies around the bias voltage; see the output waveform in Fig. 5A. Because of that, the input signal must be small enough so that its positive and negative swings do not drive the amplifier near the non-linear cutoff and saturation regions. Since a high-value resistor is used to change the output voltage to a current (I=V/R) in a class-A configuration, the output current is small. That is important since current flows at all times in such amplifiers, with or without an input signal. Power is wasted and efficiency (the ratio of output to total power consumed) is low--only about 20-25%--in call-A amplifiers. Class-A amplifiers can be configured for single-ended or push-pull operation and are used in AF (audio frequency), IF (intermediate frequency), and RF

(radio frequency) applications. Class-A operation is suitable for voltage amplifiers. In a voltage amplifier, the emphasis is on the magnitude of the output voltage. Figure 6 shows a single-ended class-A amplifier. Such an amplifier might be used in a preamplifier stage, where input signals are typically small, and a faithful reproduction of the input using a single transistor is needed. That configuration allows a small input current to control current drawn from a power source, and thus produce a stronger replica of a weaker original signal. In Class-B operation, the transistor is biased at cutoff (see Fig. 5B), so that output current flows during only half of the input cycle. It is used where high efficiency and low distortion are required--for instance, in power-output configurations. When the Class-B amplifier is used for audio applications, two such amplifiers connected in the push-pull configuration are required, so that current can flow alternately through the two amplifiers. In other words, on amplifier is turned on, while the other is turned off. On the other hand, when the Class-B amplifier is used in RF applications, it can be configured for single-ended operation. Since, in the absence of an input signal its current output is negligible, it is used where high efficiency (60-70%) and low distortion are required, which is very important in high-power amplifiers. Class-AB amplifiers (see Fig. 5C) are biased somewhere between Class-A and Class-B operation, and have efficiencies (25-35%) and distortion characteristics that lie between those of Class-A and Class-B amplifiers. Class-AB amplifiers require a somewhat larger input signal than do Class-A amplifiers. The class-AB amplifier is used in push-pull configurations for both audio-and radio-frequency applications. In Class-C operation--which has the highest efficiency (perhaps more than 90%), but offers the greatest distortion--the transistor is biased beyond the cutoff region (see Fig. 5D). Because of that, output output current flows during less than half (about a third) of the input cycle, making it unsuitable for amplifying signals of varying amplitude, such as audio. That type of amplifier is normally used to amplify a signal of fixed amplitude; for instance, it is often used in RF power output stages of a transmitter. Current in a Class-C amplifier flows in a series of power pulses that excite an LC-tank circuit into oscillation. Because of that the output waveform is a sinewave, that varies in amplitude if modulated. Class-C amplifiers can be configured for push-pull or single-ended operation. Table 1 summarizes the conduction angles and efficiency ratings of the various classes of transistor amplifier. Continue with Transistor Tutorial Part 2 Copyright © 2006 - Tony van Roon, VA3AVR Last updated: November 23, 2007

Transistors Tutorial Part 2:

"Bipolar Transistors" "The bipolar junction transistor is still one of the cornerstone's of modern solid-state electronics. Learn (or review) the basics of this important active device. The Bipolar Junction Transistor (BJT) triggered the revolution in modern solid-state electronics in the 1960's. Although the discrete small-signal BJT has since yielded to the integrated circuit (IC) in economic importance, it lives on in the form of discrete linear and switching power transistors as well as radio-frequency transistors into the microwave region. The principles behind the operation of the BJT are important to the understanding of many of today's most popular linear and digital integrated circuits. Moreover, the transistor families--TTL, Schottky TTL, and emitter-couple logic (ECL) are BJT's. This article focuses small-signal BJT's and practical circuits that can be made with them. They function either as linear amplifiers or digital switches. The term bipolar junction transistor (BJT) distinguishes it from the junction field-effect transistor or JFET.

BJT Basics: A BJT is a three-terminal (base, emitter, and collector) device. There are two types: NPN and PNP. Today both are typically made by the double-diffusion process that involves the deposition of two additional layers of doped silicon on a doped silicon wafer.

Figure 1-a shows the cross section of an NPN BJT. Its base and emitter terminals are metal depositions on top of the silicon wafer, and its collector is the metalized lower surface of the wafer. Figure 2-a shows the cross section of a PNP BJT. It is similar to the NPN BJT, except that the N- and P-type materials have changed places. Figure 1-b and 2-b are the schematic symbols for the NPN and PNP transistors, respectively. Notice that they are the same except for the direction of the arrowhead within the symbol at the emitter terminal. This difference will be explained shortly. The term bipolar means that the BJT's operation depends on the movement of two different carriers: electrons and holes. In NPN BJT's the electron is the majority carrier and the hole is the minority carrier. This situation is reversed in the PNP BJT. By contrast, all filed-effect transistors (JFET's and MOSFET's) depend upon the movement of only one carrier, either electrons or hoes, depending on whether they are Nchannel or P-channel devices, so they are technically unipolar devices. (For more on this, see Electronics Now, April and May 1993). The voltage on the collector of the NPN BJT must be positive with respect to its emitter if current Ic is to flow. That current will increase with a positive bias on the base. Figure 3-a shows how a small input current applied at the base (Ib) of the NPN BJT can control Ic. The arrowhead indicates the direction of conventional current flow--collector to emitter. Note that it is in the same direction as the arrowhead in the symbol for the NPN transistor. (Electrons flow in the direction opposing the arrowhead.) Similarly, the PNP transistor requires a negative collector supply with respect to its emitter to operate, and a negative base bias to increase conduction. Fig. 3-b shows conventional current flow in the PNP BJT from the emitter to the collector, as shown in the symbol for the NPN transistor, but opposite to that shown in Figure 3-a. Most of the common commodity NPN and PNP BJT's available from electronics distributors and retail stores have been standardized and are made by many different suppliers around the world. Table 1 lists the basic characteristics of two typical generalpurpose, small-signal BJT's that are included in the projects discussed in this article: the 2N3904 NPN-type and the 2N3906 PNP-type. Both are packaged in small, three-pin plastic cylindrical TO-92 packages with flat faces. Brief definitions for the parameters in Table 1 are:  Power dissipation is the maximum mean power that the BJT can dissipate without an external heatsink, at normal room temperature, 25°C.  FT is the gain-bandwidth product, the frequency at which the common-emitter forward current gain is unity.  VCBO is collect-base voltage (emitter open), the maximum voltage that can be impressed across collect and emitter when the base is open.  VCEO is the collector-emitter voltage (base open), the maximum voltage that can be impressed across collector and emitter when the base is open.

 IC(max) is the maximum mean current that should be allowed to flow through the collector terminal of the BJT.

 hFE is the DC forward-current gain, the ratio of DC collector current to DC base current for a transistor in a common-emitter configuration. The gain-bandwidth product, the frequency at which common emitter forward current gain is unity, applies in the following way: if a transistor in a voltage circuit has a voltage gain of X 100, its bandwidth will be one hundredth of gain bandwidth value. However if the voltage gain is reduced to X 10, the bandwidth will increase to that value divided by 10. Transistor Characteristics: A knowledge of the static and dynamic characteristics of BJT's will be useful in obtaining the optimum performance from the device. Static characteristics are values obtained when the device is in a test circuit and operated under DC conditions with the measurements made by an ohmmeter. Figure 4-a shows the static equivalent circuit of an NPN BJT, and Figure 3-b shows the static equivalent of a PNP BJT. Each device can be considered as equivalent to a pair of reverse-biased zener diodes in series between the collector and emitter terminals, with the base terminal connected to the common point between to the two zeners. Examination of Figs. 1-a and 2-a shows that each BJT is really two diodes: the emitter and base form one PN diode with an emitter-base junction, and the base and collector form a second PN diode with a base-collector junction. When these diodes are properly biased, they reach an avalanche or zener breakdown point. In most small-signal BJT's, the base-to-emitter junction has a typical zener value of 5 to 10 volts, while the base-to-collector junction has a typical zener value of 20 to 100 volts. Thus, if the base-to-emitter junction of the BJT is forward biased, it exhibits the characteristics of a zener diode. The forward-biased junction in a silicon BJT blocks virtually all current until the bias voltage rises to about 600 millivolts. Beyond that value the current will increase rapidly. When forward biased by a fixed current, the forward voltage of the junction has temperature coefficient of about -2 millivolts per degree C. When the transistor is configured as an emitter open-circuited, the base-to-collector junction exhibits similar characteristics of those just described-except for a greater zener value. If the transistor is configured with its base open-circuited, the collector-to-emitter path acts like a zener diode in series with an ordinary diode. Dynamic Characteristics: The dynamic characteristics of a BJT can be better understood by examining the typical common-emitter collector characteristics for a small-signal silicon NPN transistor shown in Fig. 5. Direct current collector current Ic is plotted on the Y axis, and DC collectoremitter voltage Vceo is plotted along the X axis. A family of curves for different values of DC base current Ib is drawn of Fig. 5. Base current is plotted because the BJT is a current-operated device. As mentioned earlier, the

understood by examining the typical common-emitter collector characteristics for a small-signal silicon NPN transistor shown in Fig. 5. Direct current collector current Ic is plotted on the Y axis, and DC collectoremitter voltage Vceo is plotted along the X axis. A family of curves for different values of DC base current Ib is drawn of Fig. 5. Base current is plotted because the BJT is a current-operated device. As mentioned earlier, the base-emitter junction is forward biased for normal transistor operation. Base current flows and is a necessary variable for establishing the BJT's operating point. Observe the following specific points on Fig. 5:  When base current (Ib is zero, the transistor conducts barely measurable collector leakage current.

 When the collector-to-emitter voltage exceeds a few hundred millivolts, the collector current value is almost directly proportional to the base current value. It is only slightly affected by the actual collector voltage value. Thus, the transistor can perform as a constant-current generator by feeding a fixed bias current into the base. The transistor can also perform as a linear amplifier by superimposing the input signal on a nominal input bias current. (This will be discussed in more detail later.) Circuit Applications: Even a simple small-signal BJT has many applications related to its ability to amplify or switch. Some of the most important and practical circuit designs are described her. With few exceptions, all of the circuits are based on the 2N3904 NPN transistor. (With certain minor component value changes, other NPN transistor can be substituted.) The circuits can also be made with a PNP transistor such as the 2N3906, if the polarities are altered. Diodes and Switches: It was explained earlier that both the base-emitter and base-collector junctions of a silicon BJT can be considered equivalent to a zener diode. As a result, either of these junctions can perform as a fast-acting rectifier diode or zener diode, depending on the bias polarity. Figure 6 shows two alternative ways to make an NPN BJT perform as a diode in a clamping circuit that converts an AC-coupled rectangular input waveform into a DC square wave. The input AC waveform is symmetrical above and below the zero-voltage reference. However, the output signal retains the input's form and amplitude, but it is clamped to the zero-voltage reference. If you build this circuit, use the base-collector terminals as the diode as in Fig. 6-b because they provide a larger zener voltage value than the circuit shown in Fig. 6-a. Figure 7 shows how an NPN BJT can function as a zener diode in a circuit that converts an unregulated supply voltage into a fixed-value regulated output voltage. Typical values range from 5 to 10 volts, depending on the characteristics of the selected transistor. The base emitter junction is the only one suitable for this application. Figure 8 shows a BJT functioning as a simple electronics switch or digital inverter. Here the base is driven through resistor Rb by a digital input step voltage that has a positive value. The load resistor Rl can be a simple resistor, tungsten lamp filament, or a relay coil. Connect the load between the collector and the positive supply. When the input voltage is zero, the transistor switch is cut off. Thus no current flows through the load, and the full supply voltage is available between the collector and emitter terminals. When the input voltage is high, the transistor switch is driven fully on. Maximum current flows in the load, and only a few hundred millivolts is developed between the collector and emitter terminals. Thus the output voltage signal is the inverted form of the input signal.

Linear Amplifiers: A BJT can function as a linear current or voltage amplifier if a suitable bias current is fed into its base, and the output signal is applied between a suitable pair of terminals. A transistor amplifier can be configured for any of three operating modes: commonemitter(Fig. 9), common-base(Fig. 10), and common-collector(Fig. 11). Each of these modes offers a unique set of characteristics. In the common-emitter circuit of Fig. 9, load resistor Rl is connected between the collector and the positive supply, and a bias current is fed into the base through Rb. The value of Rb was selected so that the collector takes on a quiescent value of about half the supply voltage (to provide maximum undistorted signal swings). The input signal in the form of a sine wave is applied between the base and the emitter through C1. The circuit inverts the phase of the input signal, which appears as an output between the collector and emitter. This circuit is characterized by a medium-value input impedance and a high overall voltage gain. The input impedance of this amplifier is between 500 and 2000 ohms, and the load impedance equals Rl. Voltage gain is the change in collector voltage divided by the change in base voltage (from 100 to about 1000). Current gain is the change in collector current divided by the change in base current of Hfe. In the common-base linear amplifier circuit of Figure 10, the base is biased through Rb and AC-decoupled (or AC-grounded) through Cb. The input signal is applied between the emitter and base through C1, and the amplified but non-inverted output signal is taken from between the collector and base. This amplifier offers very low input impedance, and output impedance equal to the resistor Rl. Voltage gain is from 100 to 1000, but current gain is near-unity. In the common-collector linear amplifier circuit of Fig. 11, the collector is connected directly to the positive voltage supply, placing it effectively at ground impedance level The input signal is applied directly between the base and ground (collector), and the noninverted output signal is taken between the emitter and ground (collector). The input impedance of this amplifier is very high; it is equal to the product of hfe and the load resistance Rl. However, output impedance is very low. The circuit's overall voltage gain is near-unity, and its output voltage is about 600 millivolts less than the input voltage. As a result, this circuit is know as a DC-voltage follower or an emitter follower. A circuit with very high input impedance can be obtained by replacing the single transistor of the amplifier of Fig. 11 with a pair of transistors connected in a Darlington configuration, as shown in Fig. 12. Here, the emitter current of the input transistor feeds directly into the base of the output transistor with an overall hfe value equal to the product of the values for the individual BJT's. For example, if each BJT has an hfe of 100, the pair acts like single transistor with an hfe of 10,000. Darlington BJT's with two transistors on a single chip (considered to be discrete device) are readily available for power amplification. The voltage-follower circuit of Fig. 11 can be modified for an alternating current input by biasing the transistor base with a value equal to half the supply voltage and feeding the

input signal to the base. Figure 14 shows how this particular circuit is structured. The emitter-follower circuits of Figs. 12 to 14 can source or feed relatively high currents into an external load through the emitter of the transistor. However, those circuits cannot sink or absorb high currents that are fed to the emitter from an external voltage source because the emitter is reverse-biased under this condition. As a result, these circuits have only a unilateral output capability. In many applications, (such as audio amplifier output stages), a bilateral output characteristic is essential. A bilateral amplifier has equal sink and source output capabilities. This is obtained with the complementary emitter-follower circuit of Fig. 14. The series-connected NPN-PNP transistor pair is biased to give a modest quiescent current through the network consisting of resistors R1 and R2 and diodes D1 and D2. Transistor Q1 can provided large source currents, and Q2 can absorb large sink currents.

Phase Splitters: Transistor linear amplifiers can be used in active filters or oscillators by connecting suitable networks between their inputs and outputs. Phase splitting is another useful linear amplifier application. It provides a pair of output signals from a single input signal: one is in phase with the input phase, and the other is inverted or 180° out of phase. Fig. 16 and 17 show these alternative circuits. In the circuit shown in Fig. 15, the BJT is connected as a common-emitter amplifier with nearly 100% negative applied through emitter resistor R4. It has the same value as collector resistor R3. This configuration provides a unity-gain inverted waveform at output 1 and a unity-gain non-inverted waveform at output 2. The phase-splitter circuit shown in Fig. 16 is known as a long-tailed pair because the two BJT's share common-emitter resistor R7. An increasing waveform applied at the base of transistor Q1 causes the voltage to increase across resistor R7, reducing the bias voltage on transistor Q2. This results in the generation of an inverted waveform at the collector of Q1 (at output 1), and an in-phase waveform at the collector of Q2, (at output 2).

Multivibrators: Figures 17 to 20 show BJT's in the four different kinds of multivibrator circuit: bistable, astable, monostable, and Schmitt trigger. The bistable multivibrator is a simple electronic circuit that has two stable states. It is more often known as the flip-flop, but is also called a binary multivibrator, or an EcclesJordan circuit. The circuit is switched from one state to the other by a pulse or other external signal. It maintains its state to the other by a pulse or other external signal. It maintains its state indefinitely unless another input signal is received. Figure 17 is a simple, manually-triggered, cross-coupled bistable multivibrator. The base bias of each transistor is obtained from the collector of the other transistor. Thus one transistor automatically turns OFF when the other turns ON, and this cycle can be continued in definitely as long as it is powered. The output of the multivibrator in Fig. 17 can be driven low by turning off transistor Q2 with switch S2. The circuit remains "locked" or stable in this state until transistor Q1 is

turned off with switch S1. At that time, the output is locked into its high state, and the process is repeated. It can be seen that this action makes it a simple digital memory circuit that holds its state until manually or electronically switched. Figure 18 is the schematic for a monostable multivibrator or one-shot pulse generator. It has only one state. The output of this circuit, a manually triggered version, is normally low, but it switches high for a period determined by the values of capacitor C1 and resistor R2 if transistor Q1 is turned off with switch S1. It then returns to tits original state. The pulse duration time of the monostable multivibrator can be determined from the equation: T = 0.69 RC Where: T is in microseconds, R is in ohms, and C is in microfarads. Monostable multivibrators are used as pulse generators and weep generators for cathoderay tubes.

Figure 19 is the schematic for an astable multivibrator or free-running, square-wave oscillator. The transistors are in a common-emitter configuration so that the output of one is fed directly to the input of the other. Two resistance-capacitor networks, R3 and C1, and R2 and C2, determine the oscillation frequency. The output of each transistor is 180° out of phase with the input. An oscillating pulse might begin at the base of Q1. It is inverted at the collector of Q1 and is sent to the base of Q2. It is again inverted at the collector of Q2 and therefore returns to the base of Q1 in its original phase. This produces positive , resulting in sustained oscillation. The astable multivibrator is frequently used as an audio oscillator, but is not usually used in radio-frequency circuits because its output is rich in harmonics. Figure 20 is a schematic for a Schmitt Trigger, a form of bistable multivibrator circuit. It produces rectangular waves, regardless of the input waveform. The circuit is widely used to convert sine waves to square waves where these is a requirement for a train of pulses with constant amplitude. The Schmitt trigger circuit remains off until the rising input waveform crosses the preset threshold trigger-voltage level set by the value of resistors R1 and R2. When transistor Q1 is switched 'on', transistor Q2 is 'off' and, the Schmitt trigger's output voltage rises abruptly. When the input signal falls back below its drop-out level, Q1 switches 'off' and Q2 switches 'on'. The output voltage of the Schmitt trigger drops to zero almost instantly. This cycle of events will then be repeated in definitely, as long as the input signal is applied. Continue with Transistor Tutorial Part 3 Copyright © 2006 - Tony van Roon, VA3AVR Last updated: December 13, 2006

Transistors Tutorial Part 3:

"Learn about common-collector bipolar junction (BJT) transistor amplifiers and apply this knowledge to the circuits that you design."

Rewritten by Tony van Roon (VA3AVR)

Common Collector Amplifier: BJT amplifiers are still widely used in modern electronic circuitry. This article focuses on practical variations of the common-collector or emitter-follower amplifier based on discrete transistors and Darlington pairs. Figure 1 shows the basic common-collector amplifier and compares it with the common-base and common-emitter amplifiers. Table 1 sums up the performance characteristics of these three bipolar amplifiers. The fundamentals of bipolar transistors were presented in Part 1 and the specifications of two widely available and typical discrete devices, the NPN 2N3904 and the PNP 2N3906 were given. The 2N3904 is included in most of the schematics in this article. The expression hfe in Table 1, known as a hybrid parameter, is the common-emitter DC forward-current gain. It is equal to the collector current divided by the base current (hfe = Ic/Ib). The value of this variable for the 2N3904 NPN transistor is typically between 100 and 300, but in this article it is considered to 200. A lot of useful information can be gained simply by studying both Fig. 1 and Table 1. THe common-collector amplifier (also widely know as the emitter-follower has its input applied between its base and collector and its output is taken across its emitter and collector. The circuit is also referred to as the grounded-collector amplifier. In practical configurations its load resistor is in series with its emitter terminal. The mathematical derivations of the results shown in Table 1 can be found in most basic texts. However, for the purposes of this article, the important characteristics of the common-collector/emitter follower amplifier to keep in mind are:

   

High input impedance Low output impedance Voltage gain approximately equal to unity Current gain approximately equal to hfe

By contrast, notice that while the common-emitter and common-base amplifiers provide high voltage gain, they offer only low-to medium input impedance. The applications for these circuits are governed by these characteristics. Digital Amplifiers: Figure 2 is the schematic for a simple NPN common-collector/emitter-follower digital amplifier. The input signal for this circuit is a pulse that swings between zero volts and the positive supply voltage. When the input of this circuit is at zero volts and the transistor is fully cut off, and the amplifier's output is also zero volts--indication zero voltage phase shift. When an input voltage exceeding +600 millivolts (the minimum forward bias for turn-on) appears across the input terminals, the transistor turns on and current IL flows in load resistor RL, generating an output voltage across RL. Inherent negative causes the output voltage to assume a value that follows the input voltage. The input voltage is equal to the input voltage minus the voltage drop across the base-emitter junction (=600 millivolt). In Fig. 2 schematic, the input (base) current is calculated as: Because the circuit can have a maximum voltage gain of one, it presents an input impedance calculated as: Inserting the values shown in Fig. 2 yields: The circuit has an output impedance that approximately equals the value of the input signal source impedance (RS) Because the circuit shown in Fig. 3 exhibits all of the common-collector amplifier characteristics previously discussed, it behaves like a unitygain buffer circuit. If high-frequency pulses are introduced at its input, the trailing edge of the output pulse will show the time constant decay curve shown in Fig. 3. This response is caused by stray capacitance CS (representing with the circuit's load resistance. When the leading edge of the input pulse switches high, Q1 switches on and rapidly sources or feeds a charge current to stray capacitance CS, thus producing and output pulse with a sharp leading edge. However, when the trailing edge of the input goes low, Q1 switches off and effective capacitor CS is unable to discharge or sink through the transistor. However, CS can discharge through lead resistor RL. That discharge will follow an exponential decay curve with the time to discharge to the 37% level equal to the product of CL and RL. Relay Drivers: The base digital or switching circuit of Fig. 2 can be put to work driving a wide variety of resistive loads such as incandescent filament lamps, LED's, or resistors. If the circuit is to drive an inductive load such as a coil, transformer, motor, or speaker, a diode much be

included to limit an input-voltage surge that could destroy the transistor when the switch is closed. The schematic in Fig. 4 is a modification of Fig. 3 with the addition of diode D1 across the load, in this case a relay coil, and switch S1 in the collector-base circuit. It can act in either the latching or non-latching modes. The relay to be actuated either by the input pulse or switch S1. Relay RY1's s close and are available for switching either when a pulse with an amplitude equal to the supply voltage is introduced or S1 is closed. The relay s open when the input pulse falls to zero or S1 is opened. Protective diode D1 damps relay RY1's switch-off voltage surge from swinging below the zero volt supply level. Optional diode D2 can also be included to prevent this voltage from rising about the positive power supply value. The addition of normally open relay 2 (RY2) makes the circuit self-latching. Figure 5 shows a same relay driver circuit organized for an PNP transistor, a 2N3906 BJT. Again, the relay can be turned on either by closing S1 or by applying the input pulse as shown. Both the circuit in Figs. 4 and 5 increase the relay's sensitivity by a factor of about 200 (Hfe value of Q1). Consider a relay requires an activating current of 100 mA and has a coil resistance of 120 ohms. The effective input impedance of the circuit (Zin) will be: 120 x 200 = 24,000 ohms. Only an input operating current of 1/200 of 100 milliamperes or 0.5 milliamperes is required. Circuit sensitivity can be further increased by replacing transistor Q1 with the Darlington pair of Q1 and Q2, as shown in Fig. 6. This circuit represents an input impedance of about 1 megohm and requires an input operating current of about 12 microamperes (uA). Capacitor C1 protects the circuit from false triggering by high-impedance transient voltages, such as those induced by lightning or electromagnetic interference. The benefits of the Darlingron pair are readily apparent in relay-driving circuits that require time delay, such as those shown in Figs. 7 and 8. In those circuits, the voltage divider formed by resistor R1 and capacitor C1 generates an waveform that rises or falls exponentially. That waveform is fed to the relay coil through the high-impedance Q1-Q2 voltagefollowing Darlington buffer. The circuit forces the relay to change state at some specified delay time after the supply voltage is applied. With the 120K resistor R1 shown in both Figs. 7 and 8, operating delays will be about 0.1 second per microfarad of capacitor value. For example, if C1 equals 100 microfarads (µF), the time delay will be 10 seconds.

In the Fig. 7 circuit, consider that C1 is fully discharged so that the R1-C1 junction is at zero volts and relay RY1 is off (s open) when the power supply is connected. Capacitor C1 then charges exponentially through R1, and the increasing voltage is fed to the relay circuit through Darlington pair Q1 and Q2. That causes relay RY1's s to close after a time delay determined by the product of R1 and C1. Consider that capacitor C1 in the Fig. 8 circuit is also fully discharged when the power supply is connected. The junction of R1 and C1 is initially at the supply voltage, and the relay close at that moment. Capacitor C1 then charges exponentially through R1, and the decaying voltage at the R1-C1 junction appears across the coil of relay RY1. The s of RY1 open after the delay determined by R1 and C1 times out.

Constant-Current Generators: A BJT can serve as a constant-current generator if it is connected in the commoncollector topology and the power supply and collector terminals function as a constantcurrent path, as shown in Fig. 9. The 1000-ohm resistor R2 is the emitter load. The series combination of resistor R1 and zener diode D1 applies a fixed 5.6-volt reference to the base of Q1.

The is a 600-millivolt (0.6V) base-to-emitter drop across Q1, so 5 volts is developed across emitter resistor R2. As a result, a fixed current of 5 milliamperes flows through this resistor from Q1's emitter. Because of a BJT's characteristics, emitter and collector currents are nearly identical. This means that a 5-milliampere current also flows in any load that is connected between Q1's collector and the circuit's positive supply. This will occur regardless of the load's resistance value--provided that the value is not so large that it drives Q1 in saturation. Therefore, these two points are constant-current source terminals. Based on the previous discussion, it can be seen that constant-current magnitude is determined by the values of the base reference voltage and emitter load resistor R2. Consequently, the value of the current can be changed by varying either of these parameters.

The Fig. 10 circuit takes this concept a step further. It can be seen, for example, that the circuit of Fig. 9 was inverted to give a ground-referenced, constant-current output. Adjustment of trimmer potentiometer R3 provides a current range of from 1 to about 10 milliamperes.

The most important feature of the constant-current circuit is its high dynamic output impedance--typically hundreds of kilo-ohms. The precise magnitude of constant current is usually unimportant in practical circuits. The circuits shown in Fig. 10 and 11 will work satisfactorily in many practical applications. If more precise current generation is required, the characteristics of the reference voltage of these circuits can be improved to eliminate the effects of power source variations and temperature changes. A simple way to improve the circuits in Figs. 9 and 10 is shown in Fig. 11. Resistor R1 in both circuits can be replaced with a 5-milliampere constant current generator. (The symbol for a constant-current generator is a pair of overlapping circles.) With a constant-current generator installed, the current through zener diode D1 and the voltage across it is independent of variations in the supply voltage. True high precision can be obtained if the industry standard reference zener diode D1 is replaced with one having a temperature coefficient of 2 millivolts/°C to match th base-toemitter temperature coefficient of transistor Q1. However, if a zener diode with those characteristics cannot be located, satisfactory results can be obtained by substituting a forward-biased light-emitting diode, as shown in Fig. 12. The voltage drop across LED1 is about 2 volts, so only about 1.4 volts appears across emitter resistor R1. If the value of R1 is reduced from 1000 to 270 ohms, the constantcurrent output level can be maintained at 5 milliamperes. Analog Amplifiers: The common-collector/emitter-follower amplifier can amplify AC-couple analog signals linearly if the transistor's base is biased to a quiescent value of about half the supply voltage. This permits maximum signal swings without distortion due to clipping. As shown in Figs. 13 and 14, the analog signals are AC-coupled to the base with capacitor C1, and the output signal is taken from the emitter through capacitor C2. Figure 13 shows the simplest analog common-collector/emitter-follower circuit. Transistor Q1 is biased by resistor R1 connected between the voltage source and the base. The value of resistor R1 must be equal to the input resistance Rin of the emitter-follower stage to obtain half-supply biasing. Input resistance Rin (and thus the nominal R1 value) equals the 4.7K value of R2 multiplied by the hFE value of the Q1 transistor. In this circuit: A slightly more elaborate biasing method is shown in Fig. 14. However, its biasing level is independent of variations in transistor Q1's hFE value. Resistors R1 and R2 function as a voltage divider that applies a quiescent half-supply voltage to Q1's base. Ideally, the value of R1 should equal the value of R2 in parallel with Rin. However, the circuit works quite well if resistor R1 has a low value with respect to Rin, and resistor R2 is slightly larger than R1. In the circuits shown in Figs. 13 and 14, the input impedance looking directly into the base of transistor Q1 equals hFE x Zload, where Zload is equal to the combined parallel impedance of R2 and any external load Zx that is connected to the output. In these circuits, the base impedance value is about 1 megohm when Zx is infinite. In practical circuits, the input impedance of the base and the bias network. The circuit shown in Fig.13 has an input impedance of about 500 kilohm, and the circuit shown in

Fig. 14 has an input impedance of about 50 kilohm. Both the Fig. 13 and 14 circuits offer a voltage gain that is slightly less than unity; the true gain is given by: Where Zb = 25/IE ohms and IE is the emitter current in milliamperes. With an operating current of 1 milliampere, these circuits provide voltage gains of 0.995 when the Zload = 4.7 kilohm, or 0.975 when the load is 1.0 kilohm. The significance of these gain figures will be discussed shortly.

Bootstrapping: The relatively low input impedance of the circuit in Fig. 14 circuit can be increased significantly by bootstrapping as illustrated in Fig. 15. The 47-kilohm resistor R3 is located between the R1-R2 junction and the base of transistor Q1, and the input signal is fed to Q1's base through capacitor C1. Notice, however, that Q1's output signal is fed back to the R1-R2 junction through C2, so that almost identical signal voltages appear at both ends of R3. Consequently, very little signal current flows in R3. The input signal "sees" far greater impedance that the true resistance value. To make this point clearer, consider that the emitter-follower circuit in Fig. 15 has a precise voltage gain of unity. In this condition, identical signal voltages would appear at the two ends of R3, so no signal current would flow in this resistor, making it "appear" equal to Rin, or 1 megohm. Practical emitter-follower circuits provide a voltage gain that is slightly less than unity. The precise gain that determines the resistor amplification factor, or AR of the circuit is: AR = 1/(1 - AV). For example, if circuit gain is 0.995 (as in Fig. 13), then AR is 200 and the R3 impedance is almost 10 megohms. By contrast, if AV = 0.975, AR is only 40 and the R3 impedance is almost 2 megohms. This impedance is effectively in parallel with Rin so, in the first example, the complete Fig. 15 circuit exhibits an input impedance of about 900 kilohm. The input impedance of the circuit in Fig. 16 circuit can be further increased by substituting a 520 Darlington pair for Q1 and increasing the value of R3, as shown in Fig. 17. This modification gives a measured input impedance of about 3.3 megohms. Alternatively, even greater input impedance can be obtained with a bootstrapped "complementary- pair" circuit as shown in Fig. 18; it offers an input impedance of about 10 megohms. In this instance, Q1 and Q2 are both connected as common-emitter amplifiers but they operate with nearly 100% negative . As a result, they provide an overall voltage gain that is almost exactly one. This transistor pair behaves like a near-perfect Darlington emitter-follower. Emitter-followers: Recall from the previous articles on bipolar transistors, a standard NPN emitter-follower can source current but cannot sink. By contrast, an PNP emitter-follower can sink current but cannot source it. This means that these circuits can only handle unidirectional output currents. A bidirectional emitter-follower (that can source or sink currents with equal ease) has many applications. This response can be obtained with a complementary emitter-follower topology--NPN and PNP emitter followers are effectively connected in series. Figures 18 to 20 illustrate some basic bidirectional emitter-follower circuits.

The circuit in Fig. 18 circuit has a dual or "split" power supply, and its output is directcoupled to a grounded load. The series connected NPN and PNP transistors are biased at a quiescent "zero volts" value through the voltage divider formed with resistors R1 and R2 and diodes D1 and D2. Each transistor is forward biased slightly with silicon diodes D1 and D2. Those diodes have characteristics that are similar to those of the transistor base-emitter junctions. Capacitor C2 assures that identical input signals are applied to each transistor base, and emitter resistors R3 and R4 protect the transistor against excessive output currents.

Transistor Q1 in Fig. 18 sources current into the load when the input goes positive, and transistor Q2 sinks load current when the input goes negative. Notice that input capacitor C1 is non-polarized. Figure 19 shows an alternative to the circuit of Fig. 18 designed for operation from a single-ended power supply and an AC-coupled output load. In this circuit, input capacitor C1 is polarized. Notice that output transistors Q1 and Q2 in Figs. 18 and 19 are slightly forward biased by silicon diodes D1 and D2 to eliminate crossover distortion problems. One diode is provided for each transistor. If these circuits are modified by substituting Darlington pairs, four biasing diodes will be required. In those variations, a single transistor "amplifier diode" stage replaces the four diodes, as shown in Fig. 20. The collector-to-emitter voltage of Q5 in Fig. 20 equals the base-to-emitter voltage drop across Q5 (about 600 millivolts, more or less) multiplied by (R3 + R4)/R4. Thus, if trimmer potentiometer R3 is set to zero ohms, about 600 millivolts are developed across Q5, which then behaves as a silicon diode. However, if R3 is set to its maximum value of 47 kilohm, about 3.6 volts is developed across Q5, which then behaves like six series connected silicon diodes. Trimmer R3 can set the voltage drop across Q5 precisely as well as adjust the quiescent current values of the Q2-Q3 stage. Continue with Transistor Tutorial Part 4: "Power Amplifiers" Copyright © 2006 - Tony van Roon, VA3AVR Last updated: September 18, 2007

Transistors Tutorial Part 4: by Tony van Roon (VA3AVR)

"Learn about audio power amplifiers and apply this knowledge to your circuits designs and experiments."

An audio power amplifier can boost weak signals from a tuner, CD player, or tape deck to fill a room with sound. This article focuses on the operating principles and circuitry of low-frequency power amplifiers based on the bipolar junction transistor (BJT). Other articles in this series have discussed multivibrators, oscillators, audio preamplifiers, and tone-control circuits, all based on the BJT. Power Amplifier Basics: A transistorized audio power amplifier converts the medium-level, medium-impedance AC signal into a high-level, amplified signal that can drive a low-impedance audio transducer such as a speaker. A properly designed power amplifier will do this with minimal signal distortion. Audio can be amplified with one or more power transistors in either of three configurations: Class A, Class B, and Class AB. Figure 1-a shows a single BJT Class A amplifier in a common-emitter configuration with a speaker as its collector load. A Class A amplifier can be identified by the way its input base is biased. Fig. 1-a shows that BJT Q1's collector current has a quiescent value that is about halfway between the zero bias and cutoff positions. (The quiescent value is that value of transistor bias at which the negative- and positive-going AC input signals are zero.) This bias permits the positive and negative swings of the output collector AC current to reach their highest values without distortion. If the AC and DC impedances of the speaker load are equal, the collector voltage will assume a quiescent value that is about half the supply voltage.

The Class A circuit amplifies audio output with minimum distortion, but transistor Q1 consumes current continuously-even in the quiescent state--giving it low efficiency. Amplifier efficiency is defined as the ratio of AC power input to the load divided by the DC power consumed by the circuit. At maximum output power, the efficiency of a typical Class A amplifier is only 40%, about 10% less than its theoretical 50% maximum. However, its efficiency falls to about 4% at one-tenth of its maximum output power level. A typical Class B amplifier is shown in Fig. 2-a. It has a pair of BJTs, Q1 and Q2, operating 180° out-of-phase driving a common output load, in this example another speaker. In this topology, the BJTs operated as common-emitter amplifiers drive the speaker through push-pull transformer T2. A phasesplitting transformer T1, provides the input drives for Q1 and Q2 180° out-of-phase. The outstanding characteristic of any Class B amplifier is that both transistors are biased off under quiescent conditions because they are operated without base bias. As a result, the amplifier draws almost no quiescent current. This gives it an efficiency that approaches 79% under all operating conditions. In Fig. 2-b, neither Q1 nor Q2 conducts until the input drive signal exceeds the base emitter zero-crossing voltage of the transistor. This occurs at about 600 millivolts for a typical power transistor. The major disadvantage of the Class B amplifier is that its output signal is seriously distorted. THis can be seen from its dynamic transfer curve, also shown in Fig. 2-b. Class AB Fundamentals: Audio distortion caused by the crossover between two out-of-phase transistors is annoying. To overcome this defect, the Class B amplifier is modified into the third category called Class AB for most high-fidelity audio equipment. Fortunately, Class B distortion can usually be eliminated by slight forward bias to the base of each transistor, as shown in Fig. 3-a. This modification sharply reduces the quiescent current of a Class B amplifier and converts it into a Class AB amplifier. Many early transistorized power amplifiers were Class AB, as shown in Fig. 3-a, but that circuit is rarely seen today. That circuit requires one transformer for input phase-splitting and another for driving the speaker, both costly electronics components. In addition, electrical characteristics of both Q1 and Q2 must be closely matched. The amplification of each transistor will be unequal if they are not, and it will be impossible to minimize output distortion. Figure 3a shows a dynamic transfer characteristic for a Class AB power amplifier. The Class AB amplifier shown in Fig. 4 avoids both transformers and the need to match transistors. A complementary pair of transistors (Q1 and NPN and Q2 a PNP) is connected as an emitter follower. Powered by a split (dual) supply, the circuit's two emitter followers are biased through R1 and R2 so that their outputs are at zero volts; no

current flows in the speaker under quiescent conditions. Nevertheless, a slight forward bias can be applied with trimmer potentiometer R3 so that Q1 and Q2 modest quiescent currents to prevent crossover distortion. Identical input signals are applied through C1 and C2 to the base of the emitter followers, which avoid a split-phase drive. When an input signal is applied to the Fig. 4 circuit, the positive swing drives PNP Q2 off while driving NPN Q1 on. Transistor Q1 acts as current source with a very low output (emitter) impedance if feeds a faithful unity-gain copy of the input voltage signal to the speaker. The transistor characteristics have little or no effect on this response. Similarly, negative swings of the input signal drive Q1 off and Q2 on. Because Q2 is a PNP BJT, it becomes a current sink with minimal input (emitter) impedance. It also produces a faithful unity-gain copy of the voltage signal to the speaker, again with Q2's characteristics having little or no effect on the circuit's response. As a result, the Fig. 4 circuit does not require that Q1 be matched to Q2, and neither input nor output transformers are required. Modification of this circuit, as shown in Figs. 5-a and b, work from single ended power supplies. In Fig. 5-a, one side of the speaker is connected to the amplifier through highvalue blocking capacitor C3 and, and the other end is connected to ground; in Fig. 5-b, one side is connected to C3 and the other side is connected to the positive supply. All three circuits are popular in modern high-fidelity audio power amplifiers based on integrated circuitry. Class AB Variations: The circuit in Figs. 4-a is a unity-voltage gain amplifier so one obvious improvement is to add a voltage-amplifying driver stage, as shown in Figs. 6. Transistor Q1, configured as a common-emitter amplifier, drives two emitter followers, Q2 and Q3, through its collector load resistor R1. Note that Q1's base bias is derived from the circuit's output through resistors R2 and R3. This configuration provides DC to stabilize the circuit's operating points and AC to minimize signal distortion. The Fig. 6 circuit illustrates how a form of auto-bias can be applied to Q2 and Q3 through the silicon diodes D1 and D2. If the simple voltage-divider biasing method in Fig. 4 is used in the Fig. 6 circuit, its quiescent current will increase as ambient temperature rises and decrease as it fall. (This is caused by the thermal characteristics of a transistor's baseemitter junction.) The biasing in Fig. 6 is derived from the forward voltage drop of series diodes D1 and D2 whose thermal characteristics are closely matched to those of the base-emitter junctions of Q2 and Q3. Consequently, this circuit offers excellent thermal compensation.

Practical amplifiers include a pre-set trimmer potentiometer in series with D1 and D2. This component makes it possible to adjust biased voltage over a limited range. Low-value resistors R4 and R5 in series with the emitters of Q2 and Q3 provide some negative DC . The impedance of the Fig. 4 circuit equals the product of the speaker load impedance and the current gain of either Q1 or Q2. The circuit can be improved by replacing transistors Q1 and Q2 with Darlingron pairs which will significantly increase the circuit's input impedance and increase the amplifier's collector load capacity. Figures 7 to 9 show three different ways of modifying the Fig. 6 circuit by replacing individual transistors with Darlington pairs. For example, in Fig. 7, transistors Q2 and Q3 form a Darlingron NPN pair, and Q4 and Q5 form a darlington PNP pair. There are four baseemitter junctions between the bases of Q2 and Q4, and the output circuit is biased with a string of four silicon diodes, D1 and D4, in series to compensate for the Darlingron pairs. Figure 8, Q2 and Q3 are a Darlington NPN pair, but Q4 and Q5 are a complementary pair of common-emitter amplifiers. They operate with 100% negative , and provide unity-voltage gain and very high input impedance. Thisquasi-complementary output stage is probably the most popular Class AB power amplifier topology today. Notice the three silicon biasing diodes, D1, D2, and D3. Finally, in Figure 9, both pairs Q2 and Q3 and Q4 and Q5 are complementary pair of unity-gain, common-emitter amplifiers with 100% negative . Because the pairs produce outputs that are mirror images of each other, the circuit has a complementary output stage. Notice that this circuit has only two silicon biasing diodes, D1 and D2. Amplified Diodes: The circuits in Figs. 6 to 9 include strings of two to four silicon biasing diodes. Each of those strings can be replaced by single transistor and two resistors configured as an amplified diode, as shown in Figs. 10. The output voltage of the circuit, Vout can be calculated from the formula: Vout = VBE x R1 + R2/R2 If resistor R1 is replaced by a short circuit, the circuit's output will be equal to the baseemitter junction "diode" voltage of Q1 (VBE). The circuit will then have the thermal characteristics of a discrete diode.

If resistor R1 equals R2, the circuit will act like two series-connected diodes, and if R1 equals three times R2, the circuit will act like four series-connected diodes, and so on. Therefore, the circuit in Figs. 10 can be made to simulate any desired whole or fractional number of series-connected diodes, depending on how the R1/R2 ratios are adjusted. Figure 11 shows how the circuit in Fig. 10 can be modified to act as a fully adjustable

"amplifier diode", with an output variable from 1 to 5.7 times the base-emitter junction voltage (VBE)

Bootstrapping: The main purpose of the Q1 driver stage in Fig. 6, the base complementary amplifier, is to give the amplifier significant voltage gain. At any given value of Q1 collector current, this voltage gain is directly proportional to the effective Q1 collector load value. It follows that the value of resistor R1 should be as large as possible to maximize voltage gain. However, there are several reasons why this does not work. First, the effective or AC value of R1 equals the actual R1 value shunted by the input impedance of the Q2-Q3 power amplifier stage. Therefore, if R1 has a higher value, the power amplifier input impedance must be even greater. That can usually be done by replacing Q2 and Q3 with high-gain transistor pairs, as was done in Figs. 7 to 9. The second reason is that Q1 in Fig. 6 must be biased so that its collector assumes a quiescent half-supply voltage value to provide maximum output signal swings; this condition is set by the Q1's collector current and resistor R1's value. The true value of R1 is predetermined by biasing requirements. To achieve high voltage gain, a way must be found to make the AC impedance of R1 much greater than its DC value. This is accomplished with he bootstrapping technique shown in Figs. 12 & 13. In Fig. 12, Q1's collector load consists of R1 and R2 in series. The circuit's output signal, which also appears across SPKR1, is fed back to the R1-R2 junction through C2. This output signal is a near unity-voltage-gain copy of the signal appearing on Q1's collector. If resistor R1 has a value of 1 kilohm, the Q2-Q3 stage provides a voltage gain of 0.9. As

a result, an undefined signal voltage appears at the low end of resistor R2, and 0.9 times that undefined voltage appears at the top of R2. In other words, only one-tenth of the unknown signal voltage is developed across R2. Therefore, it es one-tenth of the signal current that would be expected from a 1-kilohm resistor. This means that the AC signal impedance value of R2 is ten times greater (10-kilohms) than its DC value, and the signal voltage gain is increased correspondingly. In practical circuits, "bootstrapping" permits the effective voltage gain and collector load impedance of Q1 to be increased by the factor of about twenty. Fig. 13 is the schematic for an alternative version of Fig. 12 without one resistor and one capacitor. In this circuit. SPKR1 is part of Q1's collector load, and it is bootstrapped through capacitor C2. As an alternative to bootstrapping, the load resistor can be replaced with a simple transistor constant-current generator. This design is found in many integrated circuit audio power amplifiers. Alternative Drivers: Returning once again to Fig. 6, notice that parallel DC and AC voltage form the R1-R2 divider network is fed back to the Q1 driver stage. This is a simple and stable circuit, but its gain and input impedance are low. Moreover, it will work only over a limited power supply voltage range. Figure 14 is a variation of the Fig. 6 circuit intended to function as a driver stage. Current through resistors R1 and R2 allows the circuit to work over a wide supply voltage range. The resistors can be AC decoupled (as shown) through C2 to increase the gain and input impedance, but at the expense of increased signal distortion. Transistor Q1 can be replaced with a Darlington pair if very high input impedance is desired. Another alternative driver stage, Fig. 15, depends on series DC and AC to give it more gain and higher input impedance than can be obtained from the Fig. 6 circuit. In this circuit, PNP transistor Q1 is directly coupled to NPN transistor Q2. Finally, Fig. 16 is the schematic for a driver circuit specifically intended for use in amplifiers with dual or split power supplies that have directcoupled input and output stages referenced to ground. The input stage of this driver stage is a long-tailed pair. Both the input and output will be centered on DC ground if the values of resistors R1 and R4 are equal. This circuit is found in many integrated circuit

power amplifiers. An IC power amplifier: Improvements in the power-handling capabilities of monolithic integrated circuits have permitted power amplifier to be integrated on a single silicon substrate or chip. The techniques for deg integrated circuit power amplifiers are similar to those for discrete device circuits. It turns out that the similarities between discrete and IC power amplifier designs are closer than for most other linear circuits. Figure 17 is a simplified circuit diagram for the LM380, an IC power amplifier, drawn in the manufacturer's data book style. The LM380 was developed by National Semiconductor Corporation for consumer applications. It features an internally fixed gain of 50 (34 dB) and an output that automatically centers itself at one-half of the supply voltage. An unusual input stage permits inputs to be referenced to the ground or AC coupled, as required. The output stage of the LM380 is protected with both short-circuit current limiting and thermal-shutdown circuitry. The LM380 has two input terminals. Both Q1 and Q2 are connected as PNP emitter followers that drive the Q3 and Q4 differential amplifier transistor pairs. The PNP inputs reference the input to gro8und, thus permitting direct coupling of the input transducer.

The output is biased to half the supply voltage by resistor ratio R1/R2 (resistor R1 is formed by two 25-kilohm resistors and R2 has a value of 25-kilohms). Negative DC , through resistor R2, balances the differential stage with the output at half supply, because R1 = R2. The output of the differential amplifier stage is direct coupled into the base of Q12, which is a common-emitter, voltage-gain amplifier with a constant current-source load provide by Q11. Internal compensation is provided by the pole-splitting capacitor C'. Polesplitting compensation permits wide power bandwidth (100 KHz at 2 watts, 8 ohms). The collector signal of Q12 is fed to output pin 8 of the IC through the combination of

emitter-coupled Q7 and the quasi-complementary pair emitter followers Q8 and Q9. The short-circuit current is typical 1.3 amperes. Continue with Transistor Tutorial Part 5: "Audio Amplifiers"

Copyright © 2006 - Tony van Roon, VA3AVR Last updated: November 9, 2007

Transistors Tutorial Part 4: by Tony van Roon (VA3AVR)

"Learn about audio power amplifiers and apply this knowledge to your circuits designs and experiments."

An audio power amplifier can boost weak signals from a tuner, CD player, or tape deck to fill a room with sound. This article focuses on the operating principles and circuitry of low-frequency power amplifiers based on the bipolar junction transistor (BJT). Other articles in this series have discussed multivibrators, oscillators, audio preamplifiers, and tone-control circuits, all based on the BJT. Power Amplifier Basics: A transistorized audio power amplifier converts the mediumlevel, medium-impedance AC signal into a high-level, amplified signal that can drive a low-impedance audio transducer such as a speaker. A properly designed power amplifier will do this with minimal signal distortion. Audio can be amplified with one or more power transistors in either of three configurations: Class A, Class B, and Class AB. Figure 1-a shows a single BJT Class A amplifier in a common-emitter configuration with a speaker as its collector load. A Class A amplifier can be identified by the way its input base is biased. Fig. 1-a shows that BJT Q1's collector current has a quiescent value that is about halfway between the zero bias and cutoff positions. (The quiescent value is that value of transistor bias at which the negative- and positive-going AC input signals are zero.) This bias permits the positive and negative swings of the output collector AC current to reach their highest values without distortion. If the AC and DC impedances of the speaker load are equal, the collector voltage will assume a quiescent value that is about half the supply voltage. The Class A circuit amplifies audio output with minimum distortion, but transistor Q1 consumes current continuously-even in the quiescent state--giving it low efficiency. Amplifier efficiency is defined as the ratio of AC power input to the load divided by the DC power consumed by the circuit. At maximum output power, the efficiency of a typical Class A amplifier is only 40%, about 10% less than its theoretical 50% maximum. However, its efficiency falls to about

4% at one-tenth of its maximum output power level. A typical Class B amplifier is shown in Fig. 2-a. It has a pair of BJTs, Q1 and Q2, operating 180° out-of-phase driving a common output load, in this example another speaker. In this topology, the BJTs operated as common-emitter amplifiers drive the speaker through push-pull transformer T2. A phasesplitting transformer T1, provides the input drives for Q1 and Q2 180° out-of-phase. The outstanding characteristic of any Class B amplifier is that both transistors are biased off under quiescent conditions because they are operated without base bias. As a result, the amplifier draws almost no quiescent current. This gives it an efficiency that approaches 79% under all operating conditions. In Fig. 2-b, neither Q1 nor Q2 conducts until the input drive signal exceeds the base emitter zero-crossing voltage of the transistor. This occurs at about 600 millivolts for a typical power transistor. The major disadvantage of the Class B amplifier is that its output signal is seriously distorted. THis can be seen from its dynamic transfer curve, also shown in Fig. 2-b. Class AB Fundamentals: Audio distortion caused by the crossover between two out-ofphase transistors is annoying. To overcome this defect, the Class B amplifier is modified into the third category called Class AB for most high-fidelity audio equipment. Fortunately, Class B distortion can usually be eliminated by slight forward bias to the base of each transistor, as shown in Fig. 3-a. This modification sharply reduces the quiescent current of a Class B amplifier and converts it into a Class AB amplifier. Many early transistorized power amplifiers were Class AB, as shown in Fig. 3-a, but that circuit is rarely seen today. That circuit requires one transformer for input phase-splitting and another for driving the speaker, both costly electronics components. In addition, electrical characteristics of both Q1 and Q2 must be closely matched. The amplification of each transistor will be unequal if they are not, and it will be impossible to minimize output distortion. Figure 3a shows a dynamic transfer characteristic for a Class AB power amplifier. The Class AB amplifier shown in Fig. 4 avoids both transformers and the need to match transistors. A complementary pair of transistors (Q1 and NPN and Q2 a PNP) is connected as an emitter follower. Powered by a split (dual) supply, the circuit's two emitter followers are biased through R1 and R2 so that their outputs are at zero volts; no current flows in the speaker under quiescent conditions. Nevertheless, a slight forward bias can be applied with trimmer potentiometer R3 so that

Q1 and Q2 modest quiescent currents to prevent crossover distortion. Identical input signals are applied through C1 and C2 to the base of the emitter followers, which avoid a split-phase drive. When an input signal is applied to the Fig. 4 circuit, the positive swing drives PNP Q2 off while driving NPN Q1 on. Transistor Q1 acts as current source with a very low output (emitter) impedance if feeds a faithful unity-gain copy of the input voltage signal to the speaker. The transistor characteristics have little or no effect on this response. Similarly, negative swings of the input signal drive Q1 off and Q2 on. Because Q2 is a PNP BJT, it becomes a current sink with minimal input (emitter) impedance. It also produces a faithful unity-gain copy of the voltage signal to the speaker, again with Q2's characteristics having little or no effect on the circuit's response. As a result, the Fig. 4 circuit does not require that Q1 be matched to Q2, and neither input nor output transformers are required. Modification of this circuit, as shown in Figs. 5-a and b, work from single ended power supplies. In Fig. 5-a, one side of the speaker is connected to the amplifier through highvalue blocking capacitor C3 and, and the other end is connected to ground; in Fig. 5-b, one side is connected to C3 and the other side is connected to the positive supply. All three circuits are popular in modern high-fidelity audio power amplifiers based on integrated circuitry. Class AB Variations: The circuit in Figs. 4-a is a unity-voltage gain amplifier so one obvious improvement is to add a voltage-amplifying driver stage, as shown in Figs. 6. Transistor Q1, configured as a common-emitter amplifier, drives two emitter followers, Q2 and Q3, through its collector load resistor R1. Note that Q1's base bias is derived from the circuit's output through resistors R2 and R3. This configuration provides DC to stabilize the circuit's operating points and AC to minimize signal distortion. The Fig. 6 circuit illustrates how a form of auto-bias can be applied to Q2 and Q3 through the silicon diodes D1 and D2. If the simple voltage-divider biasing method in Fig. 4 is used in the Fig. 6 circuit, its quiescent current will increase as ambient temperature rises and decrease as it fall. (This is caused by the thermal characteristics of a transistor's baseemitter junction.) The biasing in Fig. 6 is derived from the forward voltage drop of series diodes D1 and D2 whose thermal characteristics are closely matched to those of the base-emitter junctions of Q2 and Q3. Consequently, this circuit offers excellent thermal compensation. Practical amplifiers include a pre-set trimmer potentiometer in series with D1 and D2. This component makes it possible to adjust biased voltage over a limited range. Lowvalue resistors R4 and R5 in series with the emitters of Q2 and Q3 provide some negative DC .

The impedance of the Fig. 4 circuit equals the product of the speaker load impedance and the current gain of either Q1 or Q2. The circuit can be improved by replacing transistors Q1 and Q2 with Darlingron pairs which will significantly increase the circuit's input impedance and increase the amplifier's collector load capacity. Figures 7 to 9 show three different ways of modifying the Fig. 6 circuit by replacing individual transistors with Darlington pairs. For example, in Fig. 7, transistors Q2 and Q3 form a Darlingron NPN pair, and Q4 and Q5 form a darlington PNP pair. There are four baseemitter junctions between the bases of Q2 and Q4, and the output circuit is biased with a string of four silicon diodes, D1 and D4, in series to compensate for the Darlingron pairs. Figure 8, Q2 and Q3 are a Darlington NPN pair, but Q4 and Q5 are a complementary pair of common-emitter amplifiers. They operate with 100% negative , and provide unity-voltage gain and very high input impedance. Thisquasi-complementary output stage is probably the most popular Class AB power amplifier topology today. Notice the three silicon biasing diodes, D1, D2, and D3. Finally, in Figure 9, both pairs Q2 and Q3 and Q4 and Q5 are complementary pair of unity-gain, common-emitter amplifiers with 100% negative . Because the pairs produce outputs that are mirror images of each other, the circuit has a complementary output stage. Notice that this circuit has only two silicon biasing diodes, D1 and D2. Amplified Diodes: The circuits in Figs. 6 to 9 include strings of two to four silicon biasing diodes. Each of those strings can be replaced by single transistor and two resistors configured as an amplified diode, as shown in Figs. 10. The output voltage of the circuit, Vout can be calculated from the formula: Vout = VBE x R1 + R2/R2 If resistor R1 is replaced by a short circuit, the circuit's output will be equal to the baseemitter junction "diode" voltage of Q1 (VBE). The circuit will then have the thermal characteristics of a discrete diode.

If resistor R1 equals R2, the circuit will act like two series-connected diodes, and if R1 equals three times R2, the circuit will act like four series-connected diodes, and so on. Therefore, the circuit in Figs. 10 can be made to simulate any desired whole or fractional number of series-connected diodes, depending on how the R1/R2 ratios are adjusted. Figure 11 shows how the circuit in Fig. 10 can be modified to act as a fully adjustable "amplifier diode", with an output variable from 1 to 5.7 times the base-emitter junction voltage (VBE)

Bootstrapping: The main purpose of the Q1 driver stage in Fig. 6, the base complementary amplifier, is to give the amplifier significant voltage gain. At any given value of Q1 collector current, this voltage gain is directly proportional to the effective Q1 collector load value. It follows that the value of resistor R1 should be as large as possible to maximize voltage gain. However, there are several reasons why this does not work. First, the effective or AC value of R1 equals the actual R1 value shunted by the input impedance of the Q2-Q3 power amplifier stage. Therefore, if R1 has a higher value, the power amplifier input impedance must be even greater. That can usually be done by replacing Q2 and Q3 with high-gain transistor pairs, as was done in Figs. 7 to 9. The second reason is that Q1 in Fig. 6 must be biased so that its collector assumes a quiescent half-supply voltage value to provide maximum output signal swings; this condition is set by the Q1's collector current and resistor R1's value. The true value of R1 is predetermined by biasing requirements. To achieve high voltage gain, a way must be found to make the AC impedance of R1 much greater than its DC value. This is accomplished with he bootstrapping technique shown in Figs. 12 & 13. In Fig. 12, Q1's collector load consists of R1 and R2 in series. The circuit's output signal, which also appears across SPKR1, is fed back to the R1-R2 junction through C2. This output signal is a near unity-voltage-gain copy of the signal appearing on Q1's collector. If resistor R1 has a value of 1 kilohm, the Q2-Q3 stage provides a voltage gain of 0.9. As a result, an undefined signal voltage appears at the low end of resistor R2, and 0.9 times that undefined voltage appears at the top of R2. In other words, only one-tenth of the unknown signal voltage is developed across R2. Therefore, it es one-tenth of the

signal current that would be expected from a 1-kilohm resistor. This means that the AC signal impedance value of R2 is ten times greater (10-kilohms) than its DC value, and the signal voltage gain is increased correspondingly. In practical circuits, "bootstrapping" permits the effective voltage gain and collector load impedance of Q1 to be increased by the factor of about twenty. Fig. 13 is the schematic for an alternative version of Fig. 12 without one resistor and one capacitor. In this circuit. SPKR1 is part of Q1's collector load, and it is bootstrapped through capacitor C2. As an alternative to bootstrapping, the load resistor can be replaced with a simple transistor constant-current generator. This design is found in many integrated circuit audio power amplifiers. Alternative Drivers: Returning once again to Fig. 6, notice that parallel DC and AC voltage form the R1-R2 divider network is fed back to the Q1 driver stage. This is a simple and stable circuit, but its gain and input impedance are low. Moreover, it will work only over a limited power supply voltage range. Figure 14 is a variation of the Fig. 6 circuit intended to function as a driver stage. Current through resistors R1 and R2 allows the circuit to work over a wide supply voltage range. The resistors can be AC decoupled (as shown) through C2 to increase the gain and input impedance, but at the expense of increased signal distortion. Transistor Q1 can be replaced with a Darlington pair if very high input impedance is desired. Another alternative driver stage, Fig. 15, depends on series DC and AC to give it more gain and higher input impedance than can be obtained from the Fig. 6 circuit. In this circuit, PNP transistor Q1 is directly coupled to NPN transistor Q2. Finally, Fig. 16 is the schematic for a driver circuit specifically intended for use in amplifiers with dual or split power supplies that have direct-coupled input and output stages referenced to ground. The input stage of this driver stage is a long-tailed pair. Both the input and output will be centered on DC ground if the values of resistors R1 and R4 are equal. This circuit is found in many integrated circuit power amplifiers. An IC power amplifier: Improvements in the power-handling capabilities of monolithic integrated circuits have permitted power amplifier to be integrated on a single silicon substrate or chip. The

techniques for deg integrated circuit power amplifiers are similar to those for discrete device circuits. It turns out that the similarities between discrete and IC power amplifier designs are closer than for most other linear circuits. Figure 17 is a simplified circuit diagram for the LM380, an IC power amplifier, drawn in the manufacturer's data book style. The LM380 was developed by National Semiconductor Corporation for consumer applications. It features an internally fixed gain of 50 (34 dB) and an output that automatically centers itself at one-half of the supply voltage. An unusual input stage permits inputs to be referenced to the ground or AC coupled, as required. The output stage of the LM380 is protected with both short-circuit current limiting and thermal-shutdown circuitry. The LM380 has two input terminals. Both Q1 and Q2 are connected as PNP emitter followers that drive the Q3 and Q4 differential amplifier transistor pairs. The PNP inputs reference the input to gro8und, thus permitting direct coupling of the input transducer.

The output is biased to half the supply voltage by resistor ratio R1/R2 (resistor R1 is formed by two 25-kilohm resistors and R2 has a value of 25-kilohms). Negative DC , through resistor R2, balances the differential stage with the output at half supply, because R1 = R2. The output of the differential amplifier stage is direct coupled into the base of Q12, which is a common-emitter, voltage-gain amplifier with a constant current-source load provide by Q11. Internal compensation is provided by the pole-splitting capacitor C'. Polesplitting compensation permits wide power bandwidth (100 KHz at 2 watts, 8 ohms). The collector signal of Q12 is fed to output pin 8 of the IC through the combination of emitter-coupled Q7 and the quasi-complementary pair emitter followers Q8 and Q9. The short-circuit current is typical 1.3 amperes. Continue with Transistor Tutorial Part 5: "Audio Amplifiers"

Copyright © 2006 - Tony van Roon, VA3AVR Last updated: November 9, 2007

Transistors Tutorial Part 4:

by Tony van Roon (VA3AVR)

"Learn about audio power amplifiers and apply this knowledge to your circuits designs and experiments."

An audio power amplifier can boost weak signals from a tuner, CD player, or tape deck to fill a room with sound. This article focuses on the operating principles and circuitry of low-frequency power amplifiers based on the bipolar junction transistor (BJT). Other articles in this series have discussed multivibrators, oscillators, audio preamplifiers, and tone-control circuits, all based on the BJT. Power Amplifier Basics: A transistorized audio power amplifier converts the mediumlevel, medium-impedance AC signal into a high-level, amplified signal that can drive a low-impedance audio transducer such as a speaker. A properly designed power amplifier will do this with minimal signal distortion. Audio can be amplified with one or more power transistors in either of three configurations: Class A, Class B, and Class AB. Figure 1-a shows a single BJT Class A amplifier in a commonemitter configuration with a speaker as its collector load. A Class A amplifier can be identified by the way its input base is biased. Fig. 1-a shows that BJT Q1's collector current has a quiescent value that is about halfway between the zero bias and cutoff positions. (The quiescent value is that value of transistor bias at which the negative- and positive-going AC input signals are zero.) This bias permits the positive and negative swings of the output collector AC current to reach their highest values without distortion. If the AC and DC impedances of the speaker load are equal, the collector voltage will assume a quiescent value that is about half the supply voltage. The Class A circuit amplifies audio output with minimum

distortion, but transistor Q1 consumes current continuously-even in the quiescent state--giving it low efficiency. Amplifier efficiency is defined as the ratio of AC power input to the load divided by the DC power consumed by the circuit. At maximum output power, the efficiency of a typical Class A amplifier is only 40%, about 10% less than its theoretical 50% maximum. However, its efficiency falls to about 4% at one-tenth of its maximum output power level. A typical Class B amplifier is shown in Fig. 2-a. It has a pair of BJTs, Q1 and Q2, operating 180° out-of-phase driving a common output load, in this example another speaker. In this topology, the BJTs operated as common-emitter amplifiers drive the speaker through push-pull transformer T2. A phasesplitting transformer T1, provides the input drives for Q1 and Q2 180° out-of-phase. The outstanding characteristic of any Class B amplifier is that both transistors are biased off under quiescent conditions because they are operated without base bias. As a result, the amplifier draws almost no quiescent current. This gives it an efficiency that approaches 79% under all operating conditions. In Fig. 2-b, neither Q1 nor Q2 conducts until the input drive signal exceeds the base emitter zero-crossing voltage of the transistor. This occurs at about 600 millivolts for a typical power transistor. The major disadvantage of the Class B amplifier is that its output signal is seriously distorted. THis can be seen from its dynamic transfer curve, also shown in Fig. 2-b. Class AB Fundamentals: Audio distortion caused by the crossover between two out-of-phase transistors is annoying. To overcome this defect, the Class B amplifier is modified into the third category called Class AB for most high-fidelity audio equipment. Fortunately, Class B distortion can usually be eliminated by slight forward bias to the base of each transistor, as shown in Fig. 3-a. This modification sharply reduces the quiescent current of a Class B amplifier and converts it into a Class AB amplifier. Many early transistorized power amplifiers were Class AB, as shown in Fig. 3-a, but that circuit is rarely seen today. That circuit requires one transformer for input phase-splitting and another for driving the speaker, both costly electronics components. In addition, electrical characteristics of both Q1 and Q2 must be closely matched. The amplification of each transistor will be unequal if they are not, and it will be impossible to minimize output distortion. Figure 3a shows a dynamic transfer characteristic for a Class AB power amplifier. The Class AB amplifier shown in Fig. 4 avoids both transformers and the need to match transistors. A complementary pair of transistors (Q1 and NPN and Q2 a PNP) is connected as an emitter follower. Powered by a split (dual) supply, the circuit's two emitter followers are biased through R1 and R2 so that their outputs are at zero volts; no current flows in the speaker under quiescent conditions.

Nevertheless, a slight forward bias can be applied with trimmer potentiometer R3 so that Q1 and Q2 modest quiescent currents to prevent crossover distortion. Identical input signals are applied through C1 and C2 to the base of the emitter followers, which avoid a split-phase drive. When an input signal is applied to the Fig. 4 circuit, the positive swing drives PNP Q2 off while driving NPN Q1 on. Transistor Q1 acts as current source with a very low output (emitter) impedance if feeds a faithful unity-gain copy of the input voltage signal to the speaker. The transistor characteristics have little or no effect on this response. Similarly, negative swings of the input signal drive Q1 off and Q2 on. Because Q2 is a PNP BJT, it becomes a current sink with minimal input (emitter) impedance. It also produces a faithful unity-gain copy of the voltage signal to the speaker, again with Q2's characteristics having little or no effect on the circuit's response. As a result, the Fig. 4 circuit does not require that Q1 be matched to Q2, and neither input nor output transformers are required. Modification of this circuit, as shown in Figs. 5-a and b, work from single ended power supplies. In Fig. 5-a, one side of the speaker is connected to the amplifier through highvalue blocking capacitor C3 and, and the other end is connected to ground; in Fig. 5-b, one side is connected to C3 and the other side is connected to the positive supply. All three circuits are popular in modern high-fidelity audio power amplifiers based on integrated circuitry. Class AB Variations: The circuit in Figs. 4-a is a unity-voltage gain amplifier so one obvious improvement is to add a voltage-amplifying driver stage, as shown in Figs. 6. Transistor Q1, configured as a common-emitter amplifier, drives two emitter followers, Q2 and Q3, through its collector load resistor R1. Note that Q1's base bias is derived from the circuit's output through resistors R2 and R3. This configuration provides DC to stabilize the circuit's operating points and AC to minimize signal distortion. The Fig. 6 circuit illustrates how a form of auto-bias can be applied to Q2 and Q3 through the silicon diodes D1 and D2. If the simple voltage-divider biasing method in Fig. 4 is used in the Fig. 6 circuit, its quiescent current will increase as ambient temperature rises and decrease as it fall. (This is caused by the thermal characteristics of a transistor's baseemitter junction.) The biasing in Fig. 6 is derived from the forward voltage drop of series diodes D1 and D2 whose thermal characteristics are closely matched to those of the base-emitter junctions of Q2 and Q3. Consequently, this circuit offers excellent thermal compensation. Practical amplifiers include a pre-set trimmer potentiometer in series with D1 and D2. This component makes it possible to adjust biased voltage over a limited range. Low-

value resistors R4 and R5 in series with the emitters of Q2 and Q3 provide some negative DC . The impedance of the Fig. 4 circuit equals the product of the speaker load impedance and the current gain of either Q1 or Q2. The circuit can be improved by replacing transistors Q1 and Q2 with Darlingron pairs which will significantly increase the circuit's input impedance and increase the amplifier's collector load capacity. Figures 7 to 9 show three different ways of modifying the Fig. 6 circuit by replacing individual transistors with Darlington pairs. For example, in Fig. 7, transistors Q2 and Q3 form a Darlingron NPN pair, and Q4 and Q5 form a darlington PNP pair. There are four baseemitter junctions between the bases of Q2 and Q4, and the output circuit is biased with a string of four silicon diodes, D1 and D4, in series to compensate for the Darlingron pairs. Figure 8, Q2 and Q3 are a Darlington NPN pair, but Q4 and Q5 are a complementary pair of common-emitter amplifiers. They operate with 100% negative , and provide unity-voltage gain and very high input impedance. Thisquasicomplementary output stage is probably the most popular Class AB power amplifier topology today. Notice the three silicon biasing diodes, D1, D2, and D3. Finally, in Figure 9, both pairs Q2 and Q3 and Q4 and Q5 are complementary pair of unity-gain, common-emitter amplifiers with 100% negative . Because the pairs produce outputs that are mirror images of each other, the circuit has a complementary output stage. Notice that this circuit has only two silicon biasing diodes, D1 and D2. Amplified Diodes: The circuits in Figs. 6 to 9 include strings of two to four silicon biasing diodes. Each of those strings can be replaced by single transistor and two resistors configured as an amplified diode, as shown in Figs. 10. The output voltage of the circuit, Vout can be calculated from the formula: Vout = VBE x R1 + R2/R2 If resistor R1 is replaced by a short circuit, the circuit's output will be equal to the baseemitter junction "diode" voltage of Q1 (VBE). The circuit will then have the thermal characteristics of a discrete diode.

If resistor R1 equals R2, the circuit will act like two series-connected diodes, and if R1 equals three times R2, the circuit will act like four series-connected diodes, and so on. Therefore, the circuit in Figs. 10 can be made to simulate any desired whole or fractional number of series-connected diodes, depending on how the R1/R2 ratios are adjusted. Figure 11 shows how the circuit in Fig. 10 can be modified to act as a fully adjustable "amplifier diode", with an output variable from 1 to 5.7 times the base-emitter junction voltage (VBE)

Bootstrapping: The main purpose of the Q1 driver stage in Fig. 6, the base complementary amplifier, is to give the amplifier significant voltage gain. At any given value of Q1 collector current, this voltage gain is directly proportional to the effective Q1 collector load value. It follows that the value of resistor R1 should be as large as possible to maximize voltage gain. However, there are several reasons why this does not work. First, the effective or AC value of R1 equals the actual R1 value shunted by the input impedance of the Q2-Q3 power amplifier stage. Therefore, if R1 has a higher value, the power amplifier input impedance must be even greater. That can usually be done by replacing Q2 and Q3 with high-gain transistor pairs, as was done in Figs. 7 to 9. The second reason is that Q1 in Fig. 6 must be biased so that its collector assumes a quiescent half-supply voltage value to provide maximum output signal swings; this condition is set by the Q1's collector current and resistor R1's value. The true value of R1 is predetermined by biasing requirements. To achieve high voltage gain, a way must be found to make the AC impedance of R1 much greater than its DC value. This is accomplished with he bootstrapping technique shown in Figs. 12 & 13. In Fig. 12, Q1's collector load consists of R1 and R2 in series. The circuit's output signal, which also appears across SPKR1, is fed back to the R1-R2 junction through C2. This output signal is a near unity-voltage-gain copy of the signal appearing on Q1's collector. If resistor R1 has a value of 1 kilohm, the Q2-Q3 stage provides a voltage gain of 0.9. As a result, an undefined signal voltage appears at the low end of resistor R2, and 0.9 times that undefined voltage appears at the top of R2. In other words, only one-tenth of the unknown signal voltage is developed across R2. Therefore, it es one-tenth of the

signal current that would be expected from a 1-kilohm resistor. This means that the AC signal impedance value of R2 is ten times greater (10-kilohms) than its DC value, and the signal voltage gain is increased correspondingly. In practical circuits, "bootstrapping" permits the effective voltage gain and collector load impedance of Q1 to be increased by the factor of about twenty. Fig. 13 is the schematic for an alternative version of Fig. 12 without one resistor and one capacitor. In this circuit. SPKR1 is part of Q1's collector load, and it is bootstrapped through capacitor C2. As an alternative to bootstrapping, the load resistor can be replaced with a simple transistor constant-current generator. This design is found in many integrated circuit audio power amplifiers. Alternative Drivers: Returning once again to Fig. 6, notice that parallel DC and AC voltage form the R1-R2 divider network is fed back to the Q1 driver stage. This is a simple and stable circuit, but its gain and input impedance are low. Moreover, it will work only over a limited power supply voltage range. Figure 14 is a variation of the Fig. 6 circuit intended to function as a driver stage. Current through resistors R1 and R2 allows the circuit to work over a wide supply voltage range. The resistors can be AC decoupled (as shown) through C2 to increase the gain and input impedance, but at the expense of increased signal distortion. Transistor Q1 can be replaced with a Darlington pair if very high input impedance is desired. Another alternative driver stage, Fig. 15, depends on series DC and AC to give it more gain and higher input impedance than can be obtained from the Fig. 6 circuit. In this circuit, PNP transistor Q1 is directly coupled to NPN transistor Q2. Finally, Fig. 16 is the schematic for a driver circuit specifically intended for use in amplifiers with dual or split power supplies that have direct-coupled input and output stages referenced to ground. The input stage of this driver stage is a long-tailed pair. Both the input and output will be centered on DC ground if the values of resistors R1 and R4 are equal. This circuit is found in many integrated circuit power amplifiers. An IC power amplifier: Improvements in the power-handling capabilities of monolithic integrated circuits have permitted power amplifier to be integrated on a single silicon substrate or chip. The

techniques for deg integrated circuit power amplifiers are similar to those for discrete device circuits. It turns out that the similarities between discrete and IC power amplifier designs are closer than for most other linear circuits. Figure 17 is a simplified circuit diagram for the LM380, an IC power amplifier, drawn in the manufacturer's data book style. The LM380 was developed by National Semiconductor Corporation for consumer applications. It features an internally fixed gain of 50 (34 dB) and an output that automatically centers itself at one-half of the supply voltage. An unusual input stage permits inputs to be referenced to the ground or AC coupled, as required. The output stage of the LM380 is protected with both short-circuit current limiting and thermal-shutdown circuitry. The LM380 has two input terminals. Both Q1 and Q2 are connected as PNP emitter followers that drive the Q3 and Q4 differential amplifier transistor pairs. The PNP inputs reference the input to gro8und, thus permitting direct coupling of the input transducer.

The output is biased to half the supply voltage by resistor ratio R1/R2 (resistor R1 is formed by two 25-kilohm resistors and R2 has a value of 25-kilohms). Negative DC , through resistor R2, balances the differential stage with the output at half supply, because R1 = R2. The output of the differential amplifier stage is direct coupled into the base of Q12, which is a common-emitter, voltage-gain amplifier with a constant current-source load provide by Q11. Internal compensation is provided by the pole-splitting capacitor C'. Polesplitting compensation permits wide power bandwidth (100 KHz at 2 watts, 8 ohms). The collector signal of Q12 is fed to output pin 8 of the IC through the combination of emitter-coupled Q7 and the quasi-complementary pair emitter followers Q8 and Q9. The short-circuit current is typical 1.3 amperes. Continue with Transistor Tutorial Part 5: "Audio Amplifiers"

Copyright © 2006 - Tony van Roon, VA3AVR Last updated: November 9, 2007

Transistors Tutorial Part 5: by Tony van Roon (VA3AVR)

"Learn about the audio amplifiers in stereos, tuners, tape/cassette, and CD players, and apply your knowledge to experiments or designs."

Transistors are the key components in many different kinds of audio preamplifiers, amplifiers, and tonecontrol circuits. Recent articles in this series have discussed the operation principles and applications for discrete bipolar junction transistors (BJT). Earlier articles have covered such subjects as low-power amplifier circuits, multivibrators, and oscillators. Audio Amplifier Basics: A modern stereo amplifier system has two closely matched high-fidelity audio amplifier channels. Typically each of those channels offers switch-selectable inputs for such signal sources as a tuner, tape-player, CD-player, TV, MTS, etc. Each also provides a single output signal to a high-power loudspeaker. To analyze one of those systems, it is useful to divide the system into three functional circuit blocks, as shown in Fig. 1. The first of these blocks is the selector/preamplifier. It permits the system listener to select the desired input signal source, and it automatically applies an appropriate amplification level and frequency correction to the signal to condition it for the second circuit block, tone/volume control. The tone/volume-control block permits the listener to adjust the frequency characteristics and the amplitude of the audible output to suit his individual taste. This block might also contain additional filter circuits including one specifically designed to screen out scratch and rumble. The last section of the amplifier system is the power amplifier. It might be able to produce power levels from a few hundred milliwatts to hundreds of watts. Audio power amplifiers are designed to cover the audio frequency range with minimal distortion. Most quality products today include automatic overload and thermal-runaway protection. The three sections of the audio amplifier system are all powered from a single built-in power supply. All three sections include individual power supply decoupling networks to prevent unwanted signal interference. The first two amplifier blocks will be discussed here. Simple Preamplifiers: The audio preamplifier circuit modifies the signal characteristics so that it will have a steady frequency response and the nominal 100-millivolt output amplitude necessary for driving the tone/volume control section. If the input signal is derived from a radio tuner or a tape player, the signal characteristics are usually in a form that can be fed directly to the tone/control section, bying the preamplifier. However, if the input is obtained from a micro-phone or other audio input device, it will probably need preamplifier conditioning. Two basic kinds of transducers are found in micro-phones and audio pickups: magnetic

or piezoelectric ceramic/crystal. Magnetic transducers typically offer low output impedance and a low signal sensitivity of about 2 millivolts. Their outputs must be fed to a highimpedance preamplifier stage with near-unity voltage gain. Most microphones have a near flat frequency response, so they can be matched to simple, flat-response preamplifier stages. Figure 2 shows a unity-gain preamplifier circuit that will work with most high-impedance ceramic or crystal microphones. It is an emitter-follower (common-collector) amplifier with an input network bootstrapped by C2 and R3. It has a typical input impedance of about 2 megohms. The combination of C5 and R5 decouples the amplifier from the DC power supply. Figures 3 and 4 show alternative preamplifier circuits that will match magnetic microphones. The single-stage circuit of Fig. 3 gives 46dB (x200) of voltage gain, and will work with most magnetic microphones. The two-stage circuit of Fig. 4, however, gives 76dB of voltage gain, and it is intended for preamplification of the output of verylow-sensitivity magnetic microphones.

RIAA Preamplifier Circuits: The replay of a constant-amplitude 20Hz to 20KHz variable-frequency signal that has been recorded on a phonograph disc with conventional stereo recording equipment will generate the nonlinear frequency response curve shown in Fig. 5. Here, the dotted line shows the idealized shape of this curve, and the solid line shows an actual shape. Examination of the idealized (dotted) version of the curve in Fig. 5 will show that the response is flat between 500 and 2120 Hz. However, it rises at a rate of 6dB/octave (20 dB/decade above 2120 Hz), and falls at a 6dB/octave rate between 500 Hz and 50 Hz. The response then flattens at frequencies below 50Hz. There are good--but difficult to explain--reasons why the precise Fig. 5 recording curves are used. However, all you really need to know is that they make it possible to produce disc recordings with excellent signal-to-noise ratios and wide dynamic ranges. The curves were applied during record pressing. The important point to be made here is that when a disc is replayed, the output of the pickup device must be ed to the power amplifier through a preamplifier whose frequency equalization curve is the mirror image (exact inverse) of the one used to make the original recording. As a result, a linear overall record-to-replay response is obtained. Figure 6 shows the RIAA equalization curve. RIAA is an abbreviation for the Recording Industry Association of America, the organization that standardized the precise specification of the curve for the equalization of phonograph records. When long-playing phonograph (record-player) records were the primary source of recorded music and audio entertainment, circuit designers had to include filter networks that corrected the input

from the record to conform to the RIAA equalization curve. The relatively recent(1994) world-wide conversion to compact discs (CDs) as the primary source of recorded music and entertainment has diminished the importance of the RIAA curve. Equalization is not required for linear signal sources such as CDs. Nevertheless, a preamplifier with an RIAA equalization network is still needed if you want to play any of the pressed long-playing and 45 rpm records. This equalization can be obtained by wiring frequency-dependent, resistive capacitive networks into a preamplifier. This circuitry causes the gain to fall as the frequency rises. One network will control the 50 to 500 Hz response, and the other will control the 2120 Hz to 20 kHz response. Figure 7 is the schematic for an amplifier with those networks that will work with any magnetic phono cartridge. It gives a 1-volt output from a 6-millivolt input at 1KHz, and provides equalization that is within 1 dB of the RIAA standard between 40 Hz and 12KHz.

The preamplifier circuit is designed around transistors Q1 and Q2, with C2 and R5, and C3 and R6 forming the resistor capacitor equalization network. The output of the emitter-follower buffer stage, transistor Q3, can be controlled by volume control potentiometer R10. The quality of reproduction of ceramic or crystal phono cartridges is generally lower than that of magnetic cartridges, but they produce far higher amplitude output signals. Ceramic and crystal phone cartridges will work with simple equalization preamplifiers-one reason why those cartridges were installed in so many low-cost record players. Figure 8 and 9 show alternative phone cartridge preamplifier/equalization circuits that can function with wither ceramic or crystal phono cartridges. Both circuits are designed around transistorized emitter-follower output stages Q1 and Q2. The output of the circuit in Fig. 8 can be controlled by volume control potentiometer R4, and that of Fig. 9 is controlled by R5. The preamplifier/equalizer in Fig. 8 will work with any phone cartridge whose capacitance is between 1000 and 10,000pF. Two-stage equalization is provided by the resistance-capacitance network made up of C1, C2, R2, and R3. Preamplification/equalization for this circuit is typically within 1.6 dB of the RIAA standard between 40 Hz and 12KHz.

The alternative preamplifier/equalizer shown in Fig. 9 will work only with phono cartridges whose capacitance value are between 5000 and 10,000pF because this capacitance is part of the circuit's frequency response network. The other part of the network is formed by C1 and R3. At 50 Hz, this circuit has a high input impedance of about 600 kilohms, which causes only slight cartridge loading. However, as frequency increases, input impedance decreases sharply, increasing cartridge loading and effectively reducing circuit gain. The equalization curve approximates the RIAA standard, and circuit performance is adequate for most practical applications. A Universal Preamplifier: Most audio amplifier systems must have preamplifiers with many different characteristics. These include high-gain linear response for magnetic microphones, lowgain linear response for tuners, and high-gain RIAA equalization for magnetic phone cartridges. To meet this broad requirement, most amplifier designers include a single universal preamplifier circuit such as the one shown in Fig. 10. Basically a high-gain linear amplifier, its characteristics can be altered by switching alternative resistor filter networks into its system. For example, when the selector switch is set to the Mag phono position, alternative input sources can be selected by S1-a, and appropriate linear-response gain control resistors R8, R9, and R10 are now selected by S1-b. Those resistor values are selected between 10 kilo ohms and 10 megohms to suit individual listener tastes. Circuit gain will be proportional, to the resistor value. Volume Control: The Volume control circuitry of an audio amplifier system is normally located between the output of the preamplifier stage and the input of the tone-control circuit. It is usually only a potentiometer within the circuit, as shown in Figs. 7, 8, and 9. However, the catch here is that rapid rotation of the potentiometer knob can apply DC voltage to the next circuit for brief intervals. That voltage could upset circuit bias and cause severe signal distortion. The block diagram in Fig. 11 shows the ideal topology and location for a volume control.

It is fully DC-isolated from the output of the preamplifier by capacitor C1, and from the input of the tone-control circuit by C2. As a result, variation of the wiper of control potentiometer R1 has no effect on the DC bias levels of either circuit. Potentiometer R1 should have a logarithmic taper, that is, its output should be logarithmic function rather than linear.

ive Tone Control: A tone-control network permits the listener to change th system amplifier's frequency response to suit his own mood or taste. He can, for example, boost or reduce the low-frequency (treble) sections of a musical selection to emphasize the sounds of specific sections of the orchestra. Tone-control networks typically consists of simple resistivecapacitive filters through which the signals are ed. Because these networks are ive, they cause some signal attenuation. Tone control networks can, if desired, be wired into the the loops of simple transistor amplifiers to give the system an overall signal gain. Those are known as active tone control circuits. Fig. 12-a shows a typical ive bass tone-control network, and Fig. 12-b through Fig. 12-d show the equivalent of this circuit when control potentiometer R3 is set to its maximum boost, maximum cut, and flat positions, respectively. Capacitors C1 and C2 are effectively open circuited when the frequency is at its lowest bass value. It can be seen from Fig. 12-b that the boost circuit is equivalent to a voltage divider formed by dividing 10 kilohms by 101 kilohms. This arrangement results in a low resistive value of about 100 ohms that only slightly attenuates bass signals. The Fig. 12-c cut circuit, by contrast, has a voltage divider equal to 100 kilohms divided by a 1 kilohm which gives a signal attenuation of about 40 dB. Finally, in Fig. 12-d when potentiometer R3 is set to the flat position, it will have 90 kilohms of resistance above the wiper and 10 kilohms below it. This circuit resistance value is equal to 100 kilohms divided by 11 kilohms. It gives a signal attenuation of about 20 dB at all frequencies. As a result, the circuit gives a maximum bass boost of about 20 dB or cut relative to the flat signals. Fig. 13 shows a typical ive treble tone-control network together with its equivalent circuits under maximum boost, maximum cut, and flat operating conditions. This circuit also provides about 20 dB of signal attenuation when potentiometer R3 is in the flat position, and it gives maximum treble boost or cut values of about 20 dB relative to its flat performance. Finally, Fig. 14 shows how the Fig. 12-a and 13-a schematics can be combined to make a complete bass and treble tone-control network The 10-kilohm resistor R5 has been added to minimize unwanted interaction between the two connected circuit sections. The input to this network can be taken from the circuit's volume control, and its output can be fed to the input of the power amplifier. Active Tone Controls: A tone-control network can be included in the path of a transistor amplifier so that the system will have an overall signal gain (rather than attenuation) when its controls are in the flat position. These networks can be simplifier versions of the basic circuit shown in Fig. 14. Fig. 15 is the schematic for an active tone-control circuit.

A comparison of Figs. 15 and 12 will reveal that the bass control section of Fig. 15 is a simplified version of Fig. 12-a. It can be seen that the two capacitors C1 and C2 of Fig. 12-a have been replaced by the single 0.039uF capacitor C2 of Fig. 15. Similarly, the treble version of Fig. 13-a, with resistors R1 and R2 eliminated. Resistors R3 and R4 balance the performance of the two section of the Fig. 15 control circuit.

An Audio Mixer: A multichannel audio mixer is an attractive modification that can be added to the volume/tone-control section of an audio amplifier. This mixer permits several different audio signals to be mixed together to form a single composite output signal. This modification will be of value if, for example, you want to hear the front-door buzzer or the sounds of a baby crying in a child's room while you listening to music. Figure 16 is the schematic for a three-channel audio mixer that will provide an overall gain of one between the output and each input channel. Each input channel includes a single 0.1 uF capacitor and a 100-kilohms resistor, to provide an output impedance of 100 kilohms. The number of input channels to this audio mixer can be increased by adding more capacitors and resistors with the same values as C1 and R1. The mixer should be located between the output of the tone-control circuitry and the input to the power amplifier. One input should be taken from the output of the tonecontrol circuit, and the other inputs should either be grounded or taken from the desired source.

Continue with Transistor Tutorial Part 6 Copyright © 2006 - Tony van Roon, VA3AVR Last updated: September 27, 2007

"Build these circuits that can amplify, filter, generate white noise, flash lamps, locate hidden metal--and perhaps even detect lies."

Rewritten by Tony van Roon

This last article on bipolar junction transistors (BJT) is a potpourri of circuits. Some are practical and some are not so practical, but they can be great for experiments. With these circuits you can amplify signals, filter high and low frequencies, generate white noise, and flash lamps. You can also boost DC voltage levels, locate hidden metal objects, and detect rising water. One circuit will even demonstrate the fundamentals of lie detection! More Power Amplifiers: Today the easiest way to build a low- to medium-power audio amplifier is to pick an integrated circuit (IC) amplifier from a manufacturer's data book and supplement it with additional components recommended in the applications notes in the data book. However, if you just want to learn amplifier principles by experimentation or you have a simple application in mind, you should build the amplifier with discrete transistors. Figure 1 is a schematic for a general-purpose, low-power, high-gain amplifier based on discrete transistors. A Class-A amplifier, it can drive a load such as a speaker or headset

with an impedance greater than 65 ohms. The amplifier draws a quiescent current of about 20 milliamperes. However, this drain can be reduced by increasing the value of R3.

Transistors Q1 and Q2 are configured as common-emitter amplifiers; the output of Q1 is directly coupled to the input of Q2. This circuit has an overall voltage gain of about 80

dB. Notice that resistor R3, the emitter load of Q2, is decoupled by capacitor C3 so that the Q2 emitter follows the average collector voltage of Q1. The base bias for Q1 is derived from Q2's emitter through R2. With this configuration, the bias is stabilized by negative DC . Input potentiometer R4 serves as the circuit's volume control. Figure 2 is the schematic for a simple, three-transistor, Class-AB complementary amplifier which can drive about 1 Watt into a 3-ohm speaker load. Transistor Q1, which is configured as a common-emitter amplifier, drives a load that is the sum of speaker SPKR1. resistor R1 and potentiometer R5. Its output voltage is followed and boosted in power by the complementary emitter-follower stage made up of Q2 and Q3. The output of the amplifier is fed through capacitor C2 to the junction of SPKR1 and R1 where it provides a low-impedance drive for SPKR1. It simultaneously bootstraps the value of R1 so that the circuit has high-voltage gain. The output is also fed back to Q1's base through R4 so that it produces a base bias through a negative loop. Carefully adjust trimmer potentiometer R5 to minimize audible signal crossover distortion so that it is consistent with lowest quiescent current consumption that can be measured. To obtain a reasonable value, set the quiescent current from 10 to 15 milliamperes. Figure 3 shows a more complex audio power amplifier that can deliver about 10 watts into a 8-ohm load when powered from a 30-volt supply. This circuit includes four, highgain, quasi-complementary output stage (Q3 to Q6). Transistor Q1 functions as an adjustable amplifier diode output biasing device in this circuit.

The main load resistor R2 of the Q2 common-emitter amplifier stage is bootstrapped by C2 and DC biased by R3. This network should set the quiescent output voltage at about half the power supply value. If it does not, alter the value of R3. The upper frequency response of the amplifier is restricted by C3, which menaces circuit stability. In addition, capacitor C5 is wired in series with R8 across the output of the amplifier to increase circuit stability. The amplifier should be set up initially as was described for the circuit in Fig. 2. Scratch/Rumble Filters: Today, with the widespread acceptance of compact discs (CDs/DVDs), records (LP) are long obsolete. However, because of the everlasting popularity of the record (it seems), the last couple years manufacturers have been bringing back the old record player in a new coat. Like myself, many people still own a large collection of these records, and when played on quality record players, they can still provide many hours of listening pleasure. Back when the records were popular, unless the record player amplifiers were properly filtered, scratch and rumble noise could interfere with reception. This interference was even more evident in the playing of the old 78 rpm records, you know, the old hard and fragile bakelite kind. While scratch and rumble are no longer universal problems, the techniques for eliminating them are still interesting. Scratch noise is essentially sound at a frequency greater than 10KHz picked up from the record's surface, while rumble is sound at a frequency typically less than 50 Hz caused by variations in turntable drive motor speed. Each of these noises can be effectively eliminated or attenuated by ing the audio output from the record player through a filter that rejects the annoying parts of the audio spectrum.

The rumble filter in Fig. 4 is a high- filter that provides unity voltage gain for all frequencies greater than 50Hz. however, it provides 12 dB per octave rejection to all frequencies below 50Hz. For example, attenuation is 40dB at 5Hz. Transistor Q1 is configured as an emitter-follower biased at about half the supply value from the lowimpedance junction formed by R1 and R2 in parallel with capacitor C3. However, negative applied through the filter network of R3, C2, C1, and R4 causes an active filter response. The rolloff frequency of the circuit can be altered, if desired, by changing the values of capacitors C1 and C2--provided that they are kept equal. For example, if the values of C1 and C2 are reduced 50% from 0.220 to 0.110 microfarads, the rolloff frequency will be double to 100Hz. The scratch filter circuit in Fig. 5 acts as a low- filter that provides unity voltage gain to all frequencies below 10KHz, but it rejects all frequencies above 10 KHz at 12dB per octave. This circuit resembles Fig. 4 except that the positions of the resistors and capacitors are transposed in the network consisting of C2, R4, C4, and R5.

The rolloff frequency of that circuit can be altered, if desired, by changing the values of C2 and C4. For example, if both are increase from 0.0022 microfarads to 0.0033microfards, the rolloff frequency is reduced from 10KHz to 7.5KHz. The circuits of Fig. 4 and 5 can be combined to make a composite scratch and rumble filter. The output of the high- filter is connected to the input of the low- filter. If desired, by switches can be installed in the individual filter sections so that the filters can easily be switched in and out of circuit. This change is illustrated schematically in Fig. 6. It's worth noting that if the circuits of Fig. 4 and 5 are built on a single board, three components can be saved by making the biasing network composed of resistors R1 and R2 and capacitor C3 common to both filter circuits. Noise Circuits: White Noise is a steady hissing sound obtained by mixing a full spectrum of randomly generated audio frequencies, each having equal sound power when averaged over time. White noise can be heard by tuning an FM radio receiver to that part of the band where no nearby station can be heard. It is intentionally generated for testing audio- and radio-frequency amplifiers. It can also be an effective sleep aid because it masks random background noises from voices, ing vehicles, car horns, closing doors, and other sources. Figure 7-a is the schematic for a simple but useful white noise generator base on the inherent white-noise generation capability of a revers-biased zener diode. In this circuit, resistor R2 and zener diode D1 form a negative- loop between the collector and base of common-emitter amplifier Q1. This loop stabilizes the DC working levels of the circuit, and capacitor C1 decouples the AC. As a result, D1 becomes a white-noise source in series with the case of Q1, which amplifiers that noise to a useful level of about 1 volt, peak-to-peak. The base-emitter junction of any silicone transistor can function as a noise-generating zener diode if its junction is reverse-biased to tits breakdown level. This breakdown typically occurs in a 2N3904 small-signal BJT at about 6 volts. Figure 7-b shows the schematic of a two-transistor, white-noise generator. In this circuit Q1 acts as a zener diode. Audio noise can be annoying, especially if you are trying to listen to a very weak broadcast station. You might find that the peaks of unwanted background noise completely swamp the broadcast signal, making it unintelligible. It is possible to overcome this problem with the noise-limiter circuit shown in Fig. 8. In this circuit, both the signal and the noise are fed to amplifier Q1 through potentiometer R3. Transistor Q1 amplifies both waveforms equally, but diodes D1 and D2 automatically limit the peak-to-peak output swing of Q1 to about 1.2 volts. If R3 is adjusted so that the

signal output is amplified to this peak level, the noise peaks will not exceed signal output. Therefore, the receiver signal will be far more intelligible. Astable Multivibrators: The astable multivibrator or square-wave generator circuit is versatile. Figure 9, for example, shows how it can flash two light-emitting diodes (Led) about once per second. Its flash rate is controlled by the time constant values of resistive-capacitive combinations of R4 and C1 and R3 and C2. The Leds are in series with the collectors of transistors Q1 and Q2, and they flash on and off symmetrically out-of-phase with each other. The flash rate can be changed by altering the values of either R4 and C1 or R3 and C2. You can also replace one of the Leds with a short circuit to make a one-Led flasher.

Figure 10 is a simple variation of the Fig. 9 astable multivibrator. This circuit generates an asymmetrical waveform at about 800 Hz, which is fed to speaker SPKR1 and limiting resistor Rx in the collector circuit of Q2. A monotone audio signal is generated when switch S1 is closed. This circuit becomes a simple sound generator if S1 is a simple on-off switch, or it can be a Morse-code practice oscillator if a telegrapher's key is substituted for S1. The frequency of the generated tone can be changed by altering the values of either or both capacitors C1 and C2. Figure 11 shows how an astable multivibrator can act as a signal injector-tracer for testing radio receivers. When S1 is in the inject position 1, transistors Q1 and Q2 are configured as a 1KHz astable multivibrator. With that setting, a sharp squarewave signal is sent to the probe terminal through R1 and C1. That waveform, which is rich in harmonics, will produce an audible output through a

radio's loudspeaker if it is injected into any audio- or radio-frequency stage of an amplitude modulated radio. By selecting a suitable injection point, the injector can help in troubleshooting a defective radio. When S1 is switched to Trace position 2, the circuit is configured as a cascaded pair of common-emitter amplifiers. The Probe input feeds the base of Q1 and Q2's output driving headphone Z1. Consequently, any weak audio signal fed to the Probe will be amplified directly and heard in the headphone. Similarly, any amplitude-modulated radio-frequency signals that are fed to the Probe will be demodulated by the non-linear response of transistor Q1, and the resulting audio signal will be amplified and heard in the earphone. If the Probe is connected at suitable test points in a radio, the tracer can troubleshoot faults.

LC Oscillators: Many applications can be found for inductancecapacitance (L/C) oscillators in test equipment and practical circuits. Figure 12 is a local oscillator Beat-Frequency Oscillator (BFO). Transistor Q1 is configured as a conventional Hartley Oscillator with modified 465 KHz Intermediate Frequency (IF) transformer as its collector load. If the internal tuning capacitor of the transformer is removed, variable capacitor C1 becomes the tuning control of a variable-frequency oscillator. The output frequency can be varied from well below 465 KHz to well above 1.7MHz. Any radio capable of receiving broadcast band frequencies will detect the oscillation frequency if it is placed near the signal generator circuit. If the signal generator is tuned to the intermediate frequency of a radio, a beat note can be heard. This will permit continuous-wave or sinus-sideband transmissions to be detected. Figure 13 is a modification of Fig. 12 without a transformer secondary. When the circuit is functioning with a nearby radio receiver acting as a detector and amplifier, it becomes a simple metal object locator. Oscillator coil L1 is made by winding 30 turns of wire tightly on a 3- to 4-inch diameter plastic core or bobbin about 1 inch long. It becomes a search head or sensing coil when it it is connected to the circuit with a 3-wire cable. The searching head or sensor can be mounted at the end of a long wooden or plastic handle if you want to use the circuit as a classic ground-sweeping metal detector. Similar circuits can detect buried treasure of military mines that include at least some metal parts. However, the complete circuit can be housed in a handheld case if you want to locate metal pipes or wiring that are hidden behind walls that are made of brick, wood, or plasterboard (gypsum/drywall). The operation of the object locator circuit in Fig. 13 depends on the presence of a metal object that will interfere with coil L1's electromagnetic field. The presence of the metal object can be detected by a battery-portable broadcast band radio held close to the locator circuit. It senses the frequency shift and gives out an audible screech. To detect a hidden metal object, first tune the radio to a local station, and then adjust C1 so that a low-frequency beat or chirp is heard from the radio's speaker. This beat note will change significantly if the locator circuit is placed near the hidden metal object. Figure 14 shows the Hartley oscillator used as a DC-to-DC converter. It is capable of converting the output of a 9-volt battery to 300-volts DC. TransformerT1 is a 9-0-9 to 250-volt transformer. Its primary forms the inductance (L) part of the oscillator. The supply voltage is stepped up to a peak of about 350 volts at T1's secondary. This waveform is rectified by half-wave rectifier diode D1, and it charges capacitor C4. With a permanent load, the output falls to about 300 volts at a load current of a few milliamperes. Caution: Without a permanent load across C4, the capacitor can deliver a powerful but non-lethal shock to the unwary! Conductive Water Switch:

A relay switch in a circuit that can be activated when a pair of probes come in with water can be very useful around the home or on a boat. It might, for example, indicate flooding of a basement, or water in the bilges of a boat. Figure 15 is a conductivewater-operated relay. Transistors Q1 and Q2 are a Darlington pair configured as a common-emitter, and relay coil RY1 is the collector load. The circuit relay is normally open (NO), but it is activated when the probes are placed across a resistance path that has a value generally less than several megohms. Most potable tap water has a bulk resistances below this value, so this circuit will work as a waterlevel relay switch. Relay RY1 can activate a pump or alarm (or both). However, the presence of salt in theater (or sea water) has higher conductivity, and it can enhance the effectiveness of the detection circuit. Because the conductivity of human shin has about the same resistance range as ordinary tap water, placing the probes in with human skin can also serve to activate the relay. Lie Detector: Figure 16 is simple Wheatstone bridge "lie-detector". However, because of the possible errors in the output of this circuit, its use should be confined to games or informal experiment. The circuit's operation is based on the knowledge that the resistance of human skin changes as a result of changes in the emotional state of the subject. The bridge of this detector circuit is formed by resistor R1 and R3 in a second arm. T4 in a third arm, and transistor a probe in its base circuit) in the fourth arm. A milliammeter with its zero point at the cent of the scale is connected across the bridge. It serves as a bridge-imbalance detector. Resistor R2 in series with the second probe is attached to the junction between Q1's collector and the low side of potentiometer R5. Large bare copper bards or silver spoons can be used to make suitable probes. The probes should be taped or strapped directly to the skin on the subject's hand or arm, separated by at least several inches. When the subject is relaxed and his or her skin resistance reaches a stable value, adjust potentiometer R5 to obtain a null on milliammeter M1. The subject can then be questioned about the truth or falsity of emotionally loaded or embarrassing subject--in fun of course. The subject's skin resistance will change in response to questions of that kind if they are phrased correctly. The bridge should be unbalanced if the subject reacts emotionally to the questions. Experiments of this kind are often performed by students taking collegelevel experimental psychology lab courses, but the equipment that they use is usually more sophisticated and sensitive than this circuit. Professional lie detectors typically factor in changes in the subject's respiration and pulse rate measured by other sensors to supplement the skin-resistance changes. The output of the machine is in the form of pen traces on a moving pater strip. Nevertheless, you might be surprised with the results you get experimenting with this simple circuit. Suggested Reading: - "The Early History of the Transistor".

- "A History of Engineering and Science in the Bell System:", Physical Sciences (19251980). S. Millman, Editor. - "Revolution in Miniature:", The History and Impact of Semiconductor electronics. - "Crystal Fire", by Michael Riordan and Lillian Hoddeson - "Transistorized!", Morgan Sparks interview. - "How we Built the Transistor" by William Shockley. New Scientist December 1972. - "The Improbable Years," Electronics (19 February 1968) - "They Had Eight Days to Learn About Transistors" Copyright and Credits: © Original author Ray Marston. Published by Gernsback Publishing. (Hugo Gernsback Publishing is (sadly) out of business since January 2000). Re-posting or taking graphics in any way or form from this website or of this project is expressly prohibited by international copyright © laws. Permission by written permission only. Continue with Transistor Tutorial Part 7 Copyright © 2006 - Tony van Roon, VA3AVR Last updated: September 26, 2007

Transistors Tutorial Part 7: "Oscillators"

"Learn about transistor oscillators and multivibrators that generate useful sine and square waves."

Rewritten and modified by Tony van Roon

Oscillators based on the bipolar junction transistor (BJT) are the subjects of this article. Previous articles in this series have included articles on the characteristics of the bipolar junction transistor, the common-collector amplifier, common-emitter and common-base voltage amplifiers, etc. Oscillator Fundamentals: An oscillator is a circuit that is capable of a sustained AC output signal obtained by converting input energy. Oscillators can be designed to generate a variety of signal waveforms, and they are convenient sources of sinusoidal AC signals for testing, control, and frequency conversion. Oscillators can also generate square waves, ramps, or pulses for switching, signalling, and control. Simple oscillators produce sinewaves, but another form, the multivibrator, produces square or sawtooth waves. These circuits were developed with vacuum-tubes, but have since been converted to transistor oscillators. Figure 1 is a simple block diagram showing an amplifier and a block representing the many oscillator phase-shift methods. Regardless of its amplifier, an oscillator must meet the two Barkhousen conditions for oscillation: 1 - The loop gain must be slightly greater than unity. 2 - The loop phase shift must be 0° or 360°. To meet these conditions the oscillator circuit must include some form of amplifier, and a portion of its output must be fed back regeneratively to the input. In other words, the voltage must be positive so it is in phase with the original excitation voltage at the input. Moreover, the must be sufficient to overcome the losses in the input circuit (gain equal to or greater than unity).

If the gain of the amplifier is less than unity, the circuit will not oscillate, and if it is significantly greater than unity, the circuit will be over-driven and produce distorted (non-sinusoidal) waveforms. As you will learn, the typical amplifier--vacuum tube, BJT, or fieldeffect transistor--imparts a 180° phase shift in the input signal, and the resistive-capacitive (RC) loop imparts the additional 180° so that the signal is returned in phase. Energy coupled back to the input by inductive methods can, however, be returned with zero phase shift with respect to the input. Specialized oscillators such as the Gunn diodes and Klystron tubes oscillate because of negative resistance effects, but the basic oscillator principles apply here as well. RC Oscillators: Figure 2 is the schematic for a phase-shift oscillator, a basic resistive-capacitive oscillator. Transistor Q1 is configured as a common-emitter amplifier, and its output (collector)signal is fed back to its input (base) through a three-stage RC ladder network, which includes R5 and C1, R2 and C2, and R3 and C3. Each of the three RC stages in this ladder introduces a 60° phase shift between its input and output terminals so the sum of those three phase shifts provides the overall 180° required for oscillation. The phase shift per stage depends on both the frequency of the input signal and the values of the resistors and capacitors in the network. The values of the three RC ladder network capacitors C1, C2, and C3 are equal as are the values of the the three resistors R5, R2, and R3. With the component values shown in Fig. 2, the 180° phase shift occurs at about 1/14 RC or 700 Hz. Because the transistor shifts the phase of the incoming signal 180°, the circuit also oscillates at about 700 Hz. At the oscillation frequency, the three-stage ladder network has an attenuation factor of about 29. The gain of the transistor can be adjusted with trimmer potentiometer R6 in the emitter circuit to compensate for signal loss and provide the near unity gain required for generating stable sinewaves. To ensure stable oscillation, R6 should be set to obtain a slightly distorted sinewave output. The amplitude of the output signal can be varied with trimmer potentiometer R4. Although this simple phase-shift oscillator requires only a single transistor, it has several drawbacks: poor gain stability and limited tuning range. There are ways to overcome the drawbacks of the phase-shift oscillator, and one of them is to include a Wien-bridge or network in the oscillator's loop. The concept is illustrated in the Fig. 3 block diagram. A far more versatile RC oscillator than the phaseshift oscillator, its operating frequency can be varied easily.

As shown within the dotted box in Fig. 3, a Wien Bridge consists of a series-connected resistor and capacitor, wired to a parallel- connected resistor and capacitor. The component values are "balanced" so that R1 equals R2 and C1 equals C2. The Wien Network is exceptionally sensitive to frequency. That shift is negative (to a maximum of -90°) at low frequencies, and positive (to a maximum of +90°) at high frequencies. It is zero a center frequency of 1/6.28RC. At the center frequency, network attenuation is a factor of 3. As a result, the Wien network will oscillate if a non-inverting, amplifier with a gain of 3 is connected as shown between the amplifier's output and input terminals. The output is taken between the output of the amplifier and ground. A basic two-stage Wien-Bridge oscillator schematic is shown in Fig. 4. Both transistors Q1 and Q2 are configured as low-gain common-emitter amplifiers. The voltage gain of Q2 is slightly greater than unity, and it provides the 180° phase shift required for regenerative . The 4.7K resistor R4, part of the Wien bridge network, functions as the oscillator's collector load. Transistor Q1 provides the high input impedance for the output of the Wien network. Trimmer potentiometer R5 will set the oscillator's gain over a limited range. With the component values shown, the Wien bridge oscillator will oscillate at about 1 KHz. Trimmer R5 should be adjusted so that the sinewave output signal is just slightly distorted to achieve its maximum stability. Many different practical variable-frequency Wien-bridge oscillators can be built with operational amplifier integrated circuits combined with an automatic gain-control network. No inductors are needed in these circuits.

LC Oscillators: Resistive-capacitive sine wave oscillators can generate signals from a few hertz up to several megahertz, but inductive-capacitive (LC) oscillators can generate sinewave outputs from 20 or 30 KHz up to UHF frequencies. An LC oscillator includes an LC network that provides the frequencyselective between the output of the amplifier and its input terminals. Because of the inherently high Q or frequency selectivity of LC networks or resonant tank circuits, LC oscillators produce more precise sinewave outputs--even when the loop gain of the circuit is far greater than unity.

The tuned-collector oscillator shown in Fig. 5 is the simplest of many different LC oscillators. Transistor Q1 is configured as a commonemitter amplifier, with its base bias provided by the junction of series resistors R1 and R2. Emitter resistor R3 is decoupled from highfrequency signals by capacitor C3. The primary turns of transformer T1 (L1) in parallel with trimmer capacitor C1 form a tuned collector resonant tank circuit. Collector-tobase is provided by coil L2 in transformer T1. Coil L2, with a smaller number of turns than L1, is inductively couple to L1 by transformer action. The necessary zero phase shift around the loop can be obtained by adjusting trimmer capacitor C1. If loop gain exceeds unity at the tuned frequency, the circuit will oscillate. Loop gain is determined by the turns ratio of L1 with respect to L2 in transformer T1. The phase relationship between the energizing current of all LC tuned circuits and inducted voltage varies over the range of -90° to +90°, and it is zero at a center frequency given by the formula: . Because the circuit in Fig. 5 provides a 0° overall phase shift, it oscillates at this center frequency. The frequency can be varied by trimmer capacitor C1 from 1MHz to 2 MHz. The circuit can be enhanced to oscillate at frequencies from less than 100 Hz to UHF (Ultra High Frequency) frequencies with a laminated iron-core transformer. The same circuit will oscillate satisfactorily in the UHF regions with an air-core transformer. Classic LC Oscillators: Figure 6 illustrates the Hartley Oscillator, which is a variation of the tuned-collector oscillator that was shown in Fig. 5. This oscillator is recognizable by the tapped coil in its tuned resonant circuit. Oscillation of the Hartley oscillator circuit depends on phasesplitting autotransformer action of the tapped coil in the tuned resonant circuit. The tap is located on load inductor L1 about 20% of the way down from its top so that about 1/5 of the turns are above the tap and 4/5 are below. The positive power supply is connected to the tap to obtain the necessary autotransformer action. The signal voltage across the top of L1 is 180° out-of-phase with he signal voltage across its lower end, which is connected to the collector of Q1. The signal from the top of the coil is coupled to the base (input) of Q1 through isolating capacitor C2. The oscillator will oscillate at a center frequency determined by its LC product. The Colpitts Oscillator shown in Fig. 7 is another classic circuit. It is identified by the voltage divider in its tuned resonant circuit. With the component values shown, this Colpitts circuit will oscillate at about 37KHz. Capacitor C1 is in parallel with the output capacitance of Q1, and C2 is in parallel with the input capacitance of Q1. Consequently, changes in Q1's capacitance (due to

temperature changes or aging) can shift the oscillator frequency. This shift can be minimized for high frequency stability by selecting values of C1 and C2 that are relative to the internal capacitances of Q1. The Clapp Oscillator, a modification of the Colpitts oscillator, shown in Fig. 8, offers higher frequency stability than the Colpitts oscillator. This is achieved by adding capacitor C1 in series with the coil in the tuned resonant tank circuit. It is selected to have a value that is small with respect to C2 and C3. As a result of the presence of this capacitor, the resonant frequency of the tank and oscillator will be determined primarily by the values of L1 and C1. Capacitor C3 essentially eliminates transistor capacitance variations as a factor in determining the Clapp oscillator's resonant frequency. With the component values shown, the Clapp oscillator oscillates at about 80KHz. Figure 9 shows the classic Reinartz Oscillator. In this circuit, tuning coil L1 in the collector circuit and the tuning coil L2 in the emitter circuit are inductively coupled to tuning coil L3 in the resonant tank circuit. Positive is obtained by coupling the collector and emitter signals of the transistor through windings L1 and L2, and inductively coupling both of these coils to L3. This Reinartz oscillator oscillates at a frequency determined by L3 and trimmer capacitor C2. With the values and turns ratios given in Fig. 9, the circuit will oscillate at a few hundred KHz. Modulation: The LC oscillator circuits shown in Figs. 5 to 9 can be modified to produce amplitude- or frequencymodulated (AM or FM) rather than continuous wave (CW) output signals. Figure 10 is the schematic for a beat-frequency oscillator (BFO). It is based on the tunedcollector circuit of Fig. 5, but modified to become a 465-KHz amplitude-modulated (AM) BFO. A standard 465-KHz IF transformer (T1), intended for transistor circuits, is the LC resonant tank circuit in this oscillator. An audio-frequency AM signal fed to the emitter of Q1 through blocking capacitor C2 will modulate the supply voltage of Q1 and thus amplitude-modulate the circuit's 465-KHz carrier signal.

This BFO can provide 40% signal modulation. The value of emitterdecoupling capacitor C1 was selected to present a low impedance to the 465-KHz carrier signal, while also presenting a high impedance to the low-frequency modulation signal.

Figure 11 shows how the BFO circuit in Fig. 10 can be modified to become a frequency modulator. Tuning is adjusted by trimmer potentiometer R5. Silicon diode D1 functions as an inexpensive varactor diode. A 1N4001 diode frequency modulates the 465-KHz BFO circuit. Here, C2 and diode "capacitor" D1 are in series. Consequently, the oscillator's center frequency cam be changed by altering the capacitance of D1 with trimmer potentiometer R5, and frequency-modulated signals can be obtained by introducing an audiofrequency modulation signal to D1 through C1 and R4. Capacitor C2 provides DC isolation between Q1 and D1. Astable Oscillators: Conventional oscillator circuits produce sinewaves, but repetitive square waves are important in electronics. One way to generate them is with the astable multivibrator circuit shown in Fig. 12a. This multivibrator is a self-oscillating regenerative switch whose on and off periods are controlled by the time constants obtained as the products of R2 and C2, and R3 and C1. If these time constants are equal (because both values of R and C are equal), the circuit becomes a square-wave generator that operates at a frequency of about 1/1.4 RC. The waveforms taken at the collector and base of transistors Q1 and Q2 are shown in Fig. 12b. The frequency of the astable multivibrator in Fig. 12 can be decreased by increasing the values of C1 and C2 or R2 and R3, or increased by decreasing them. The frequency can be varied with dual-gang variable resistors placed in series with 10K limiting resistors in place of R2 and R3. The operating frequency can, if required, be synchronized to that of a higher-frequency signal by coupling part of the external signal into the timing networks of the astable circuit. Outputs can be taken from either collector of the circuit, and the two outputs are in opposite in phase. The multivibrator's operating frequency is essentially independent of power supply voltage between + 1.5 and + 9 volts. The upper voltage limit is set by inherent transistor behavior: as the transistors change state at the end of each half-cycle, the base-emitter junction of one transistor is reverse biased by a voltage that is about equal to the supply voltage. Consequently, if the supply voltage exceeds the reverse base-emitter breakdown voltage of the transistor, circuit timing will be affected. This characteristic can be overcome with the circuitry modifications shown in Fig. 13. A silicon diode is connected in series with the base input terminal of each transistor to raise the effective base-emitter reverse breakdown voltage of each transistor to a value greater than that of the diode.

The protected astable multivibrator will operate with any supply voltage from +3 to +20 volts. Its frequency will vary only about 2% when the supply voltage is varied from +6 to +18 volts. This variation can be further reduced to 0.5% by adding another "compensation" diode in series with the collector of each transistor, as shown in Fig. 13. Multivibrator Variations: The basic astable multivibrator shown in Fig. 12 can be modified in different ways to improve its performance or change the shape of its output waveform. Some modifications are shown in Figs. 14 to 18. A shortcoming of the multivibrator shown in Fig. 12 is that the leading edge of each of its output waveforms is slightly rounded. The lower the values of timing resistors R2 and R3 with respect to collector load resistors R1 and R4, the more pronounced will be this waveform rounding. Conversely, the larger the values of R2 and R3 with respect to R1 and R4, the sharper the waveform edge will be. The maximum permissible values of R1 and R4 are, however, limited by the current gains of the transistors. These gains are equal to hFE multiplied wither by the value of resistor R1 or R4. One way to improve the circuit waveform, of course, would be to replace transistors Q1 and Q2 with Darlington transistor pairs and then substitute timing resistance values that are as large as permissible. That is done in the long-period astable multivibrator that is shown in Fig. 14. Resistors R2 and R3 can have any value between 10K and 12Meg, and the multivibrator will run from any supply voltage between +3 and +18 volts. With R2 and R3 values shown in Fig. 14, the multivibrator's total period or cycle time is about 1 second per microfarad when C1 and C2 have equal values. This multivibrator generates sharpcornered square waves. The square waves with the rounded leading edges produced by the multivibrator shown in Fig. 12 are caused by an inherent characteristic of the transistor. As each transistor is switched off, its collector voltage is prevented from switching abruptly to the positive supply value. This is due to the loading between that collector and the base of the adjacent conducting transistor from timing capacitor cross-coupling. This characteristic can be altered, and sharp square waves can be obtained by effectively disconnecting the timing capacitor from the collector of its transistor as it turns off. That improvement is shown in Fig. 15, a schematic for a 1-KHz astable multivibrator. It includes diodes D1 and D2 that disconnect the timing capacitors at the moment of switching. The important time constants of the multivibrator in Fig. 15 are also determined by C1 and R4, and C2 and R1. The effective collector loads of Q1 and Q2 are equal to the parallel resistances of R1 and R2, and R5 and R6, respectively. Basic astable multivibrator operation depends on slight differences in their transistor characteristics. Those differences cause one transistor to turn on faster than the other when power is first applied, thus triggering oscillation.

If the multivibrator's supply voltage is applied slowly by increasing it from zero, however, both transistors could turn on simultaneously. If this happens, the oscillator will be a nonstarter. The possibility of nonstarting can be avoided with the "sure-start" astable multivibrator circuit shown in Fig. 16. There, the timing resistors are connected to the transistor's collectors so that only one transistor can conduct at a time, ensuring that oscillation will always begin. All the astable multivibrator circuits shown so far are intended to produce symmetrical output waveforms, with a 1 to 1 ratio of square wave to space (1:1 mark/space ratio). Non-symmetrical waveforms can be obtained by installing one set of RC astable time constant components that is larger than the other. Figure 17 shows a 1.1KHz variable mark/space ratio generator. The ratio can be varied over the range 1 to 10 with trimmer potentiometer R5. However, the leading edges of the output waveforms of this circuit could be too round for some applications when mark/space control is set at its extreme position. Also, this generator could be difficult to start if the supply voltage is applied slowly to the circuit. Both of the drawbacks can be overcome with the modifications shown in the schematic of Fig. 18, another 1.1KHz variable mark/space ratio generator. The circuit includes both sure-start and waveform-correction diodes. Copyright and Credits: © 1993 Original author Ray Marston. Electronics Now, December 1993. Published by Gernsback Publishing. (Hugo Gernsback Publishing is (sadly) out of business since January 2000). All graphics, drawings, photos, © 2006 Tony van Roon. Re-posting or taking graphics in any way or form from this website or of this project is expressly prohibited by international copyright laws. Permission by written consent only.

Continue with Transistor Tutorial Part 8 Copyright © 2006 - Tony van Roon, VA3AVR Last updated: October 3, 2007

Transistors Tutorial Part 7: "Oscillators"

"Learn about transistor oscillators and multivibrators that generate useful sine and square waves."

Rewritten and modified by Tony van Roon

Oscillators based on the bipolar junction transistor (BJT) are the subjects of this article. Previous articles in this series have included articles on the characteristics of the bipolar junction transistor, the common-collector amplifier, common-emitter and common-base voltage amplifiers, etc. Oscillator Fundamentals: An oscillator is a circuit that is capable of a sustained AC output signal obtained by converting input energy. Oscillators can be designed to generate a variety of signal waveforms, and they are convenient sources of sinusoidal AC signals for testing, control, and frequency conversion. Oscillators can also generate square waves, ramps, or pulses for switching, signalling, and control. Simple oscillators produce sinewaves, but another form, the multivibrator, produces square or sawtooth waves. These circuits were developed with vacuum-tubes, but have since been converted to transistor oscillators. Figure 1 is a simple block diagram showing an amplifier and a block representing the many oscillator phase-shift methods. Regardless of its amplifier, an oscillator must meet the two Barkhousen conditions for oscillation: 1 - The loop gain must be slightly greater than unity. 2 - The loop phase shift must be 0° or 360°. To meet these conditions the oscillator circuit must include some form of amplifier, and a portion of its output must be fed back regeneratively to the input. In other words, the voltage must be positive so it is in phase with the original excitation voltage at the input. Moreover, the must be sufficient to overcome the losses in the input circuit (gain equal to or greater than unity).

If the gain of the amplifier is less than unity, the circuit will not oscillate, and if it is significantly greater than unity, the circuit will be over-driven and produce distorted (non-sinusoidal) waveforms. As you will learn, the typical amplifier--vacuum tube, BJT, or fieldeffect transistor--imparts a 180° phase shift in the input signal, and the resistive-capacitive (RC) loop imparts the additional 180° so that the signal is returned in phase. Energy coupled back to the input by inductive methods can, however, be returned with zero phase shift with respect to the input. Specialized oscillators such as the Gunn diodes and Klystron tubes oscillate because of negative resistance effects, but the basic oscillator principles apply here as well. RC Oscillators: Figure 2 is the schematic for a phase-shift oscillator, a basic resistive-capacitive oscillator. Transistor Q1 is configured as a common-emitter amplifier, and its output (collector)signal is fed back to its input (base) through a three-stage RC ladder network, which includes R5 and C1, R2 and C2, and R3 and C3. Each of the three RC stages in this ladder introduces a 60° phase shift between its input and output terminals so the sum of those three phase shifts provides the overall 180° required for oscillation. The phase shift per stage depends on both the frequency of the input signal and the values of the resistors and capacitors in the network. The values of the three RC ladder network capacitors C1, C2, and C3 are equal as are the values of the the three resistors R5, R2, and R3. With the component values shown in Fig. 2, the 180° phase shift occurs at about 1/14 RC or 700 Hz. Because the transistor shifts the phase of the incoming signal 180°, the circuit also oscillates at about 700 Hz. At the oscillation frequency, the three-stage ladder network has an attenuation factor of about 29. The gain of the transistor can be adjusted with trimmer potentiometer R6 in the emitter circuit to compensate for signal loss and provide the near unity gain required for generating stable sinewaves. To ensure stable oscillation, R6 should be set to obtain a slightly distorted sinewave output. The amplitude of the output signal can be varied with trimmer potentiometer R4. Although this simple phase-shift oscillator requires only a single transistor, it has several drawbacks: poor gain stability and limited tuning range. There are ways to overcome the drawbacks of the phase-shift oscillator, and one of them is to include a Wien-bridge or network in the oscillator's loop. The concept is illustrated in the Fig. 3 block diagram. A far more versatile RC oscillator than the phaseshift oscillator, its operating frequency can be varied easily.

As shown within the dotted box in Fig. 3, a Wien Bridge consists of a series-connected resistor and capacitor, wired to a parallel- connected resistor and capacitor. The component values are "balanced" so that R1 equals R2 and C1 equals C2. The Wien Network is exceptionally sensitive to frequency. That shift is negative (to a maximum of -90°) at low frequencies, and positive (to a maximum of +90°) at high frequencies. It is zero a center frequency of 1/6.28RC. At the center frequency, network attenuation is a factor of 3. As a result, the Wien network will oscillate if a non-inverting, amplifier with a gain of 3 is connected as shown between the amplifier's output and input terminals. The output is taken between the output of the amplifier and ground. A basic two-stage Wien-Bridge oscillator schematic is shown in Fig. 4. Both transistors Q1 and Q2 are configured as low-gain common-emitter amplifiers. The voltage gain of Q2 is slightly greater than unity, and it provides the 180° phase shift required for regenerative . The 4.7K resistor R4, part of the Wien bridge network, functions as the oscillator's collector load. Transistor Q1 provides the high input impedance for the output of the Wien network. Trimmer potentiometer R5 will set the oscillator's gain over a limited range. With the component values shown, the Wien bridge oscillator will oscillate at about 1 KHz. Trimmer R5 should be adjusted so that the sinewave output signal is just slightly distorted to achieve its maximum stability. Many different practical variable-frequency Wien-bridge oscillators can be built with operational amplifier integrated circuits combined with an automatic gain-control network. No inductors are needed in these circuits.

LC Oscillators: Resistive-capacitive sine wave oscillators can generate signals from a few hertz up to several megahertz, but inductive-capacitive (LC) oscillators can generate sinewave outputs from 20 or 30 KHz up to UHF frequencies. An LC oscillator includes an LC network that provides the frequencyselective between the output of the amplifier and its input terminals. Because of the inherently high Q or frequency selectivity of LC networks or resonant tank circuits, LC oscillators produce more precise sinewave outputs--even when the loop gain of the circuit is far greater than unity.

The tuned-collector oscillator shown in Fig. 5 is the simplest of many different LC oscillators. Transistor Q1 is configured as a commonemitter amplifier, with its base bias provided by the junction of series resistors R1 and R2. Emitter resistor R3 is decoupled from highfrequency signals by capacitor C3. The primary turns of transformer T1 (L1) in parallel with trimmer capacitor C1 form a tuned collector resonant tank circuit. Collector-tobase is provided by coil L2 in transformer T1. Coil L2, with a smaller number of turns than L1, is inductively couple to L1 by transformer action. The necessary zero phase shift around the loop can be obtained by adjusting trimmer capacitor C1. If loop gain exceeds unity at the tuned frequency, the circuit will oscillate. Loop gain is determined by the turns ratio of L1 with respect to L2 in transformer T1. The phase relationship between the energizing current of all LC tuned circuits and inducted voltage varies over the range of -90° to +90°, and it is zero at a center frequency given by the formula: . Because the circuit in Fig. 5 provides a 0° overall phase shift, it oscillates at this center frequency. The frequency can be varied by trimmer capacitor C1 from 1MHz to 2 MHz. The circuit can be enhanced to oscillate at frequencies from less than 100 Hz to UHF (Ultra High Frequency) frequencies with a laminated iron-core transformer. The same circuit will oscillate satisfactorily in the UHF regions with an air-core transformer. Classic LC Oscillators: Figure 6 illustrates the Hartley Oscillator, which is a variation of the tuned-collector oscillator that was shown in Fig. 5. This oscillator is recognizable by the tapped coil in its tuned resonant circuit. Oscillation of the Hartley oscillator circuit depends on phasesplitting autotransformer action of the tapped coil in the tuned resonant circuit. The tap is located on load inductor L1 about 20% of the way down from its top so that about 1/5 of the turns are above the tap and 4/5 are below. The positive power supply is connected to the tap to obtain the necessary autotransformer action. The signal voltage across the top of L1 is 180° out-of-phase with he signal voltage across its lower end, which is connected to the collector of Q1. The signal from the top of the coil is coupled to the base (input) of Q1 through isolating capacitor C2. The oscillator will oscillate at a center frequency determined by its LC product. The Colpitts Oscillator shown in Fig. 7 is another classic circuit. It is identified by the voltage divider in its tuned resonant circuit. With the component values shown, this Colpitts circuit will oscillate at about 37KHz. Capacitor C1 is in parallel with the output capacitance of Q1, and C2 is in parallel with the input capacitance of Q1. Consequently, changes in Q1's capacitance (due to

temperature changes or aging) can shift the oscillator frequency. This shift can be minimized for high frequency stability by selecting values of C1 and C2 that are relative to the internal capacitances of Q1. The Clapp Oscillator, a modification of the Colpitts oscillator, shown in Fig. 8, offers higher frequency stability than the Colpitts oscillator. This is achieved by adding capacitor C1 in series with the coil in the tuned resonant tank circuit. It is selected to have a value that is small with respect to C2 and C3. As a result of the presence of this capacitor, the resonant frequency of the tank and oscillator will be determined primarily by the values of L1 and C1. Capacitor C3 essentially eliminates transistor capacitance variations as a factor in determining the Clapp oscillator's resonant frequency. With the component values shown, the Clapp oscillator oscillates at about 80KHz. Figure 9 shows the classic Reinartz Oscillator. In this circuit, tuning coil L1 in the collector circuit and the tuning coil L2 in the emitter circuit are inductively coupled to tuning coil L3 in the resonant tank circuit. Positive is obtained by coupling the collector and emitter signals of the transistor through windings L1 and L2, and inductively coupling both of these coils to L3. This Reinartz oscillator oscillates at a frequency determined by L3 and trimmer capacitor C2. With the values and turns ratios given in Fig. 9, the circuit will oscillate at a few hundred KHz. Modulation: The LC oscillator circuits shown in Figs. 5 to 9 can be modified to produce amplitude- or frequencymodulated (AM or FM) rather than continuous wave (CW) output signals. Figure 10 is the schematic for a beat-frequency oscillator (BFO). It is based on the tunedcollector circuit of Fig. 5, but modified to become a 465-KHz amplitude-modulated (AM) BFO. A standard 465-KHz IF transformer (T1), intended for transistor circuits, is the LC resonant tank circuit in this oscillator. An audio-frequency AM signal fed to the emitter of Q1 through blocking capacitor C2 will modulate the supply voltage of Q1 and thus amplitude-modulate the circuit's 465-KHz carrier signal.

This BFO can provide 40% signal modulation. The value of emitterdecoupling capacitor C1 was selected to present a low impedance to the 465-KHz carrier signal, while also presenting a high impedance to the low-frequency modulation signal.

Figure 11 shows how the BFO circuit in Fig. 10 can be modified to become a frequency modulator. Tuning is adjusted by trimmer potentiometer R5. Silicon diode D1 functions as an inexpensive varactor diode. A 1N4001 diode frequency modulates the 465-KHz BFO circuit. Here, C2 and diode "capacitor" D1 are in series. Consequently, the oscillator's center frequency cam be changed by altering the capacitance of D1 with trimmer potentiometer R5, and frequency-modulated signals can be obtained by introducing an audiofrequency modulation signal to D1 through C1 and R4. Capacitor C2 provides DC isolation between Q1 and D1. Astable Oscillators: Conventional oscillator circuits produce sinewaves, but repetitive square waves are important in electronics. One way to generate them is with the astable multivibrator circuit shown in Fig. 12a. This multivibrator is a self-oscillating regenerative switch whose on and off periods are controlled by the time constants obtained as the products of R2 and C2, and R3 and C1. If these time constants are equal (because both values of R and C are equal), the circuit becomes a square-wave generator that operates at a frequency of about 1/1.4 RC. The waveforms taken at the collector and base of transistors Q1 and Q2 are shown in Fig. 12b. The frequency of the astable multivibrator in Fig. 12 can be decreased by increasing the values of C1 and C2 or R2 and R3, or increased by decreasing them. The frequency can be varied with dual-gang variable resistors placed in series with 10K limiting resistors in place of R2 and R3. The operating frequency can, if required, be synchronized to that of a higher-frequency signal by coupling part of the external signal into the timing networks of the astable circuit. Outputs can be taken from either collector of the circuit, and the two outputs are in opposite in phase. The multivibrator's operating frequency is essentially independent of power supply voltage between + 1.5 and + 9 volts. The upper voltage limit is set by inherent transistor behavior: as the transistors change state at the end of each half-cycle, the base-emitter junction of one transistor is reverse biased by a voltage that is about equal to the supply voltage. Consequently, if the supply voltage exceeds the reverse base-emitter breakdown voltage of the transistor, circuit timing will be affected. This characteristic can be overcome with the circuitry modifications shown in Fig. 13. A silicon diode is connected in series with the base input terminal of each transistor to raise the effective base-emitter reverse breakdown voltage of each transistor to a value greater than that of the diode.

The protected astable multivibrator will operate with any supply voltage from +3 to +20 volts. Its frequency will vary only about 2% when the supply voltage is varied from +6 to +18 volts. This variation can be further reduced to 0.5% by adding another "compensation" diode in series with the collector of each transistor, as shown in Fig. 13. Multivibrator Variations: The basic astable multivibrator shown in Fig. 12 can be modified in different ways to improve its performance or change the shape of its output waveform. Some modifications are shown in Figs. 14 to 18. A shortcoming of the multivibrator shown in Fig. 12 is that the leading edge of each of its output waveforms is slightly rounded. The lower the values of timing resistors R2 and R3 with respect to collector load resistors R1 and R4, the more pronounced will be this waveform rounding. Conversely, the larger the values of R2 and R3 with respect to R1 and R4, the sharper the waveform edge will be. The maximum permissible values of R1 and R4 are, however, limited by the current gains of the transistors. These gains are equal to hFE multiplied wither by the value of resistor R1 or R4. One way to improve the circuit waveform, of course, would be to replace transistors Q1 and Q2 with Darlington transistor pairs and then substitute timing resistance values that are as large as permissible. That is done in the long-period astable multivibrator that is shown in Fig. 14. Resistors R2 and R3 can have any value between 10K and 12Meg, and the multivibrator will run from any supply voltage between +3 and +18 volts. With R2 and R3 values shown in Fig. 14, the multivibrator's total period or cycle time is about 1 second per microfarad when C1 and C2 have equal values. This multivibrator generates sharpcornered square waves. The square waves with the rounded leading edges produced by the multivibrator shown in Fig. 12 are caused by an inherent characteristic of the transistor. As each transistor is switched off, its collector voltage is prevented from switching abruptly to the positive supply value. This is due to the loading between that collector and the base of the adjacent conducting transistor from timing capacitor cross-coupling. This characteristic can be altered, and sharp square waves can be obtained by effectively disconnecting the timing capacitor from the collector of its transistor as it turns off. That improvement is shown in Fig. 15, a schematic for a 1-KHz astable multivibrator. It includes diodes D1 and D2 that disconnect the timing capacitors at the moment of switching. The important time constants of the multivibrator in Fig. 15 are also determined by C1 and R4, and C2 and R1. The effective collector loads of Q1 and Q2 are equal to the parallel resistances of R1 and R2, and R5 and R6, respectively. Basic astable multivibrator operation depends on slight differences in their transistor characteristics. Those differences cause one transistor to turn on faster than the other when power is first applied, thus triggering oscillation.

If the multivibrator's supply voltage is applied slowly by increasing it from zero, however, both transistors could turn on simultaneously. If this happens, the oscillator will be a nonstarter. The possibility of nonstarting can be avoided with the "sure-start" astable multivibrator circuit shown in Fig. 16. There, the timing resistors are connected to the transistor's collectors so that only one transistor can conduct at a time, ensuring that oscillation will always begin. All the astable multivibrator circuits shown so far are intended to produce symmetrical output waveforms, with a 1 to 1 ratio of square wave to space (1:1 mark/space ratio). Non-symmetrical waveforms can be obtained by installing one set of RC astable time constant components that is larger than the other. Figure 17 shows a 1.1KHz variable mark/space ratio generator. The ratio can be varied over the range 1 to 10 with trimmer potentiometer R5. However, the leading edges of the output waveforms of this circuit could be too round for some applications when mark/space control is set at its extreme position. Also, this generator could be difficult to start if the supply voltage is applied slowly to the circuit. Both of the drawbacks can be overcome with the modifications shown in the schematic of Fig. 18, another 1.1KHz variable mark/space ratio generator. The circuit includes both sure-start and waveform-correction diodes. Copyright and Credits: © 1993 Original author Ray Marston. Electronics Now, December 1993. Published by Gernsback Publishing. (Hugo Gernsback Publishing is (sadly) out of business since January 2000). All graphics, drawings, photos, © 2006 Tony van Roon. Re-posting or taking graphics in any way or form from this website or of this project is expressly prohibited by international copyright laws. Permission by written consent only.

Continue with Transistor Tutorial Part 8 Copyright © 2006 - Tony van Roon, VA3AVR Last updated: October 3, 2007

Transistors Tutorial Part 8: "Amplifier Design"

"It's easy to design a simple transistor amplifier. Here is how."

Rewritten & modified by Tony van Roon We were having trouble finding an exact replacement transistor while repairing a piece of equipment. Figuring that an exact replacement was going to be impossible to find, we began to discuss what to do. And someone pointed out that there were only two kinds of bipolar transistors--NPN and PNP. Of course, values for various characteristics vary widely, even for a specific transistor; but in many circuits, a garden-variety device will work (and did in our case). Deg and repairing transistorized circuits is much simple than you might suspect. A well-designed circuit has built-in tolerance, so it's probably not device sensitive. The most important characteristics to consider when substituting devices or deg a circuit from scratch are operating frequency and power level. What follows is the design procedure we went through to solve an audio-gain problem.

Try it when you need a little extra gain for that next audio project. An Audio Amp: This particular project involved injecting the audio from a TV receiver into a stereo system. (These days even the cheapest TV has that feature, including MTS stereo inputs for digital accessories). Anyways, the audio-output portion of the TV-audio receiver was abandoned because of its poor frequency response and high distortion. Instead, we wanted to come right off the detector into a quality audio amplifier and speaker. So, after picking off the audio at a convenient point in the set (in this case, from a potentiometer), we wanted to feed it to the auxiliary input of the stereo amplifier. The amplifier we used required an input of 1 volt RMS, but a quick check with an AC VTVM indicated that out picked-off audio signal was only 0.1-volt RMS. Obviously, an amplifier with a gain of 10 was needed. Scanning the literature on transistor amplifiers reveled that a common-emitter amplifier with a voltage-divider bias circuit would solve our problem nicely. Such a circuit is shown in Fig. 1. Some of that circuit's characteristics include: moderate input impedance, moderate voltage gain, inverted output, and input/output impedance and gain that depend only slight on transistor beta. There are, of course, several rules that must be followed in using a common-emitter amplifier, including:  With a positive supply use an NPN transistor.  With a negative supply use an PNP transistor.  The supply voltage must not exceed the transistor's Vce rating.  The power-dissipation rating of the transistor must not be exceeded.  The beta of the transistor should be 100 or higher. In our example the following facts are known:  Our amplifier had a single-ended 12-volt power supply.  We need a voltage gain of 100.  The input impedance of the amplifier should be about 15K, the same as the potentiometer from which the audio was taken.  The impedance of the stereo amplifier's auxiliary input is about 50K. As is the case in most circuit designs, a few facts are known, and the rest must be calculated or picked using a a few "rules of thumb". We will learn how to make the calculations next. Doing the Math: For maximum undistorted output swing, we will make the quiescent collector voltage 1/2 the supply voltage. See Fig. 2. The drop across Rc must therefore be 6 volts. The value of Rc, the collector load resistance, is chosen considering output impedance, gain, and collector current. If possible, the output impedance should be lower than the

impedance of the circuit we are feeding by a factor of 10 or more. Doing so will avoid circuit loading. So let's make Rc equal to 4700 ohms, which is about 50K/10. Collector current Ic, is equal to 0.5Vcc/Rc, or 6/4700 = 1.28 mA. That current is certainly low enough that we will not exceed any collector-current ratings, so let's go on. To achieve maximum stability, the emitter resistor should be in the range of 40 to 1000 ohms. Voltage gain (Av) = Rc/Re, so Re = Rc/Av. In our case Re equals 4700/10, or 470 ohms. That falls within the range of acceptable values. The current through the emitter resistor consists of the collector current plus the base current. The base current here is significantly smaller than the collector current, so it can be ignored for the next calculation. The voltage drop across the emitter resistor = Ic X Re, or 1.28 mA x 470 ohms = 0.602 volts. The base voltage must exceed the emitter voltage by 0.6 volts for a silicon transistor and by 0.2 volts for a germanium transistor. We'll use a silicon transistor (most if not all germanium types are obsolete) in our circuit, so the base voltage must be 0.6 + 0.602 = 1.202 volts. The input impedance of the circuit equals R2 in parallel with the emitter resistor times beta; input impedance will vary with the transistor's beta. FOr our example, assume we are using a transistor with a beta of 100. We want the input impedance to be about 15000 ohms. Solving for R2, we find: Zin R2 R2 R2

= = = =

(R2 X Re X beta)/[R2 + (Re X beta)] (Zin X Re X beta)[(Re X beta) - Zin] (15000 x 470 x 100)/[470 x 100) - 15000] 22,030 ohms.

We can use a 22K resistor. In general, if input impedance is not critical, for maximum stability R2 can be 10 to 20 times Re. The drop across R2 must be 1.20 volts so the current through R2 is 1.20/22,000, or 0.054 mA. Therefore, R1 must drop the rest of the supply voltage, which is 12 - 1.20 = 10.8 volts. The current flowing through R1 is a combination of the voltage-divider current plus the base current. The base current is equal to the collector current divided by beta. It is found from: Ibeta = 1.28/100 = 0.0128 mA

So the total current through R1 is 0.054mA + 0.0128mA = 0.067mA, and R1 = 10.8/0.067mA = 160,000 ohms (160K). Resistor R1 is the most critical resistor in the circuit. To ensure maximum voltage swing, it should bring the quiescent collector voltage to one half the supply voltage. After building the circuit, the value of R1 may have to be varied slightly to achieve that voltage swing. We now have a circuit we can test. Interfacing: Connecting the circuit to the outside world will require capacitor coupling. That serves to isolate the AC signal from any DC bias voltages. Figure 3 shows our complete circuit with input and output coupling capacitors. The values of those capacitors were calculated

using C = 1/(3.2 x ƒ x R), where C equals the capacitor value in farads, ƒ equals the frequency at which response will be down 1dB, and R equals the impedance on the load side of the capacitor. To calculate the value of C1, the amplifier's input impedance (15K) is used for R. To calculate the value of C2, the input impedance of the next stage (50K) is used for R. The value of C1 can now be calculated for a drop of 1dB at 20 Hz: C1 = 1/(3.2 x 20 x 15000) = .00000104 farad = 1.0 uF. The value of C2 = 1/(3.2 x 20 x 50000) = .00000031 farad = 0.33uF. To increase the gain of the stage, you could by Re with a capacitor, as shown in Fig. 4. Nothing comes for free, however. The price you pay for increase gain is lower input impedance, which will vary widely with beta. If that variation is not a problem, a significant gain increase can be realized by adding the by capacitor. Our original circuit has a gain of 10; if the emitter is byed the gain becomes Rc/003/Ie = 4700/ (0.03/0.00129) = 4700/23 = 200 (approx). The value of the by capacitor in farads is calculated from the formula C = 1/(6.2 x ƒ x R). Again ƒ is the low-frequency limit in Hz, and R is the dynamic emitter resistance (0.031/Ie). In our example, if we stick to a 20-Hz lower limit we have C = 1/[6.2 x 20 x (0.03/0.00129)] = .000344 farads = 344 uF. A 350uF unit can be used. Thoughts: A few thoughts on components before we finish: using 5% resistors allows closer adherence to the calculated values. Because of their temperature stability and low leakage specifications, silicon rather than germanium transistors are preferable for this type of circuit. Finally, you've no doubt noticed that we have yet to specify a specific transistor. That's because for this type of application it really doesn't matter! Almost any small signal device will do fine. Copyright and Credits: Copyright © of the original article by author Jack Cunkelman, published in Radio Electronics Magazine, August 1987. Published by Gernsback Publishing. (Hugo Gernsback Publishing is (sadly) out of business since January 2000). Re-posting or taking graphics in any way or form from this website or of this project is expressly prohibited by international copyright © laws. Permission by written request only. Continue with Transistor Tutorial Part 9, FET's Copyright © 2006 - Tony van Roon, VA3AVR Last updated: November 5, 2007

Transistors Tutorial Part 9: "Learn about field-effect transistor: JFET's, MOSFET's, DMOS MOSFET's, and CMOS-how they are made, how they work, and what to look for when selecting them for your designs."

Rewritten and modified by Tony van Roon

Field Effect Transistors (FET's): Field Effect Transistors (FET's) are unipolar rather than bipolar devices, and this gives them certain properties that are superior to those of bipolar transistors. Unlike the bipolar transistor, whose current depends on the movement of both electrons and holes, FET operation depends on only one of those charge carriers. Freed from the time delays that occur when those charge carrier recombine, FET's offer faster switching sped and higher cutoff frequencies. Other advantages of the FET include:  Voltage rather than current operation.  Extremely high input impedance in the 'OFF' state.  Virtually constant current with respect to voltage at specific bias levels.  Current change that is inversely rather tan directly proportional to temperature. Despite these advantages, the FET has not replace the bipolar transistor in all applications, but it has encouraged new generations of small-signal and general purpose and RF MOSFET's as well as general purpose and RF MOSFET power transistors. Moreover, the latest digital logic families are based on FET technologies. The basic FET is a simple, three-terminal, voltage-controlled device with characteristics that are similar to those of vacuum-tube pentodes. Thus, the FET is considered to be the solid-state equivalent of a pentode. The two major classes of FET's are the Junction FET (JFET) and Metal-OxideSemiconductor FET (MOSFET), formerly called an insulated-gate-FET (IGFET). FET's are further divided into N-channel, P-channel, depletion-mode, and enhancement-mode devices. MOSFET's with N-doped channels are called NMOS, and those with P-doped channels are referred to as PMOS. The three electrodes in all FET's are the source, drain and gate, analogous to the emitter, collector, and base of the bipolar transistor. Both JFET's and MOSFET's are available as discrete transistors. Some MOSFET's have dual gates and are intended for use as radiofrequency mixers. This article explains how P-channel adn N-channel MOSTFET's are combined to form the popular complementary-MOS (CMOS) digital logic families. CMOS technology has made possible very large scale integrated (VLSI) memories, microprocessors and dedicated circuits that contain more than one million transistors and still have very low power requirements. Most small-signal FET's today are fabricated with planar geometry in which all electrodes are accessible from the top of the device. The active regions are defined on the wafer or substrate by successive masking, etching, and deposition or ion-implantation

steps. But power MOSFET' now being fabricated with vertical structures are actually capable of handling much higher current in smaller areas of silicon. Junction FET's: The simplest FET, the JFET, is illustrated by the cross-section view of Fig. 1-a. It is made by selectively implanting or diffusing ions into the wafer or substrate. An N-type region is defined on the P-type substrate by photolithographic methods, and N-type ions are implanted to form the Nchannel. Later in the manufacturing process, following further masking, oxide-deposition and etching steps, P-type ions are implanted or diffused into the N-channel to form the Ptype gate. Aluminum source and drain terminal are formed directly on the N-channel and an aluminum gate terminal is formed to the P-type gate. The symmetrical construction of the JFET permits the drain adn source to be interchanged, if necessary. If a positive voltage is applied at the drain of the N-channel JFET shown in Fig. 1-a, and a negative voltage is applied at the source with the gate terminal open, a drain current flows. When the gate is biased negative with respect to the source, the PN junction is reverse biased, and a depletion region, devoid of current carriers, is formed. Because the N-channel is more lightly doped than the P-type gate material, the depletion region penetrates into the N-channel. This region, depleted of charge carriers, behaves like an insulator. The depletion region narrows the N-channel and increases its resistance. If the bias is made even more negative, drain current is cut off completely. The gate-bias voltage that cuts off the drain current is called the pinchoff or gate-cut-off voltage. However, as the bias becomes positive, the depletion region recedes, the channel resistance is reduced, and drain current increases. Thus, the JFET gate actually controls JFET current. The schematic symbol for the N-channel JFET is shown in Fig. 1-b. As in other schematic symbols for solid-state devices, the arrowhead (representing the direction of conventional current flow) points form P-doped material to N-doped material. In the N-channel JFET symbol, the arrowhead points from the P-type gate toward the N-type channel. A section view of a P-channel JFET is shown in Fig. 2-a. The channel of the device is Ptype material, and the gate is N-type. If a positive voltage is applied to the source, conventional current flows from the source to the drain. To reverse bias the junction between the N-type gate and the P-type channel, the gate must be made positive with

respect to the channel. The biasing voltages of a P-channel JFET are opposite to those of the N-channel JFET> The schematic symbol for the P-channel JFET is shown in Fig. 2-b. The arrowhead also points form P-type material to N-type material. In this instance, it points from the P-type channel to the N-type gate region. The characteristics of the P-channel JFET are similar to those of the N-channel device, except that the voltage and current polarities are reversed. Both N-type and P-type JFET's operate in the depletion mode; that is, they conduct with zero bias on their gates. Figure 3 shows a typical family of drain characteristics for an Nchannel JFET. As the gate-to-source voltage is made increasingly negative, the depletion region is increased, and drain current decreases. As a result, pinchoff voltage occurs at a lower value of VDS. Curves for different values of gate-to-source bias, VGS, are plotted in the figure because the FET is a voltage-operated device. JFET Circuits: When an N-channel JFET is connected to a VDS supply as shown in Fig. 4, a drain current, ID can be controlled by a gate-to-source bias voltage, VDS. Similarly, when a P-channel JFET is connected to a negative drain voltage, a drain current, ID, flows in the device. The value of ID is maximum when VGS equals zero, and it is reduced (to bring the JFET into a linear operating region) by applying a reverse bias to the gate terminal of the device (negative bias in a N-channel devices, positive bias in a P-type). In Fig. 3, the value of VGS to reduce ID to zero, the gate-to-source pinchoff voltage VP is about -7 volts. The value of ID when VGS equals zero (called IDSS or saturation current for zero bias) is about 52 milli-amperes for the device shown in the figure. The gate-to-source junction of the JFET has the characteristics of a silicon diode. When reverse biases (to bring it into its linear operating region), gate leakage currents (IGSS) are measured in thousandths of a micro-ampere at room temperature. Actual gate signal currents are only a fr4action of that, and the input impedance to the gate is typically 1000 megohms at low frequencies. The gate junction is effectively shunted by a capacitance of a few picofarads, so input impedance falls as input frequency is increased. If the gate-to-source junction of the JFET is forward biased, it conducts like a normal silicon diode, and if it is severely reverse biased it avalanches like a Zener diode. Neither of those conditions will harm a JFET if its gate currents are limited to those specified. Referring to the N-channel JFET drain characteristics in Fig. 3, it can be seen that, for each value of VGS, drain current ID rises linearly from zero as the drain-to-source voltage (VDS) is increases from zero to a value at which a knee occurs on each curve. Moreover, ID remains virtually constant as VDS is increased beyond where the knee occurs.

Thus, when VDS for any of the family of VGS curves is below its knee value, the drain-to-source pins of the JFET act like a voltage-variable resistor with value determined by VGS. The drain-to-source resistance, RDS, can be varied from several hundred ohms at VGS = zero to thousands of megohms at pinchoff. That characteristic permits the JFET to be used in a circuit as a voltage-controlled switch. From the drain characteristic curve of Fig. 3, it can be seen that when VDS is above the knee value, the ID value is dictated primarily by the VGS value, and is virtually independent of the VDS value. This characteristic permits the JFET to function as a voltage-controlled current generator. The gain of a JFET is specified as a transconductance, gm, the rate of change of drain current with respect to gate voltage. A gm of 5 milli-amperes per volt indicates that a variation of one volt on the gate produces a change of 5 milli-amperes ID. The units of this measurement are in inverse ohms or mhos. You will find that JFET data sheets usually specify gm in millimhos or micromhos. The N-channel JFET in Fig. 4 is organized as a common-source amplifier, analogous to a bipolar NPN common-emitter amplifier. In typical applications, the JFET is biased into its linear region and organized as a voltage-to-voltage converter or amplifier. As shown in Fig. 4, a load resistor of suitable value, RL, should be placed in series with the JFET's drain-to-source current. Another common JFET configuration is the common drain or source-follower configuration shown in Fig. 5. That configuration is analogous to the bipolar emitterfollower configuration. Yet another possible JFET configuration is the common-gate configuration shown in Figure 6. That configuration is analogous to a bipolar commonbase configuration. MOSFET's Explained: The metal-oxide-FET or MOSFET was developed as an improvement on the JFET, and it has become the most important form of FET. Figure 7-a illustrates an N-channel depletion-mode MOSFET with a negative gate bias. The gate of this MOSFET is fully insulated from the adjacent channel. This is the most important distinction between an Ntype depletion-mode MOSFET and an N-type JFET, which is manufactured with a doped gate region directly under and in with the gate. The surface of the silicon P-type wafer is first coated with a layer of silicon dioxide (SiO2), and the source and drain windows are masked and etched to expose the P-type substrate. N-dopants are heavily diffused or implanted into those regions. Another window is masked and etched over the channel, and it is given a lighter concentration of N dopant. In subsequent steps, the channel is recoated with an insulating oxide, and the metal source, drain, and gate terminals are deposited.

When the drain is positive with respect to the source, a drain current will flow, even with zero gate voltage. However, if the gate is made negative with respect to the substrate, positive charge carriers (holes) induced in the N-channel will combine with the electrons and cause channel resistance to increase. With increasing negative bias, the pinchoff voltage will be reached, and drain current will cease. However, if the gate is made positive with respect to the substrate, additional electrons are induced, and the channel current then increases. The schematic symbol for the N-type depletion-mode MOSFET is shown in Figure 7-b. The path or channel between the source and drain is shown as a solid bar. THe symbol for the P-channel depletion-mode MOSFET is identical to the N-type, except that the arrow points outwards. Figure 8 is a drain-to-source characteristic curve for an N-channel depletionmode MOSFET. It can be seen that the current drain, ID, is inversely proportional to the magnitude of the negative gate voltages, VGS. Compare Fig. 8 with Fig. 3 for the Nchannel JFET to see their similarities. Planar enhancement-mode MOSFET's are made by the same methods as planar depletion-mode MOSFET's. However, the N-channel enhancement-mode MOSFET shown in Fig. 9 does not have the N-doped drain-to-source channel through the P-type substrate of the N-channel depletion-mode MOSFET. Therefore, there is no conduction between drain and source at zero gate bias. To turn an enhancement-mode MOSFET on, positive gate bias is needed. As the gate voltage is increased, more electrons are induced into the channel. They cannot flow across the oxide layer of the gate, so they accumulate at the substrate surface below the gate oxide. When a sufficient number of electrons has accumulated, the P-type substrate material is converted into an N-channel, and drain-to-source conduction occurs. The magnitude of the drain current depends on the channel resistance, but it is controlled by the gate voltage. The schematic symbol for an N-type enhancement-mode MOSFET is shown in Fig. 9-b. In this symbol, the gate does not make direct with the channel. The arrowhead points from the P-type substrate toward the (induced) N-type channel, shown as a line broken into three sections to indicate an intermittent current channel. Current flow in the channels of both kinds of enhancement-mode MOSFET's is proportional to the voltage on their gates, VGS. this can be seen for an N-type enhancement-mode MOSFET by examining the family of gate voltage VGS curves in Fig. 10. Current drain, ID, is directly proportional to the positive value of gate voltage.

A P-channel enhancement-mode MOSFET is made the same way as the N-channel device except that P-channel drain and source regions are diffused into an N-type substrate. The symbol for a P-type enhancement-mode MOSFET is the same as the one shown in Fig. 9-b except that the direction of the arrow is reversed. In the case of a P-type enhancement-mode MOSFET, the drain current is directly proportional to the negative values of its grid voltage. The high gate impedance of all MOSFET's, makes them susceptible to damage from even low-energy electrostatic discharge (ESD). For this reason many discrete MOSFET's and IC's based on MOSFET's are protected with on-chip Zener diode circuits. CMOS Logic Devices: An enhancement-mode MOSFET can act as a switch when it is turned on or off by a voltage applied to the gate electrode: N-channel MOSFET's are switched with positive gate voltage, and P-channel MOSFET's are switched with negative voltage. These are known as complementary responses, and they form the basis for complementary MOS or CMOS digital logic families. Figure 11-a is a section view of a complementary pair of MOSFET's on a common substrate, the basic topography for all CMOS gates. The common substrate that is used for this pair is an N-doped silicon wafer. To make an Nchannel MOSFET on an N-doped substrate, it is necessary to diffuse or implant a Pdoped well in the substrate. The smaller N-type wells can then be formed in this P-doped region. Because the substrate is N-doped, fewer steps are required to form the P-channel FET. The P- and N-doped guard bands isolate and insulate the individual transistors in this integrated circuit to prevent mutual interference. Although not illustrated here, these guard bands are actually N- or P-doped rings formed around the complete FET below the oxide layer in this CMOS technology. The two transistors in the section view, Fig. 11-a, can be connected to form a CMOS logic inverter, the simplest of digital logic circuits. This is accomplished by connecting the gates together to form an input (VIN) terminal, and taking the output (VOUT) from the common drain. The source on the left side of the diagram, VSS, is grounded, while the source on the right side is connected to the positive supply, VDD. Those connections are schematically shown in Figure 11-b. How does the inverter work? Consider the P-channel device to be the driver and the N-channel device to be the load. Recall that an N-channel enhancement MOSFET conducts with a negative gate voltage.

When the voltage input to the inverter is low (logic 0), the gate voltage of the P-channel device is negative, equal to the supply voltage VDD. As a result, the P-channel MOSFET is switched on, and there is a low impedance path from the output to VDD. Because the Nchannel is off (gate voltage is zero), there is a very high impedance path from the output to ground. Therefore, the output voltage rises to VDD. When the input voltage is high (logic 1), the situation is reversed. The P-channel FET is cut off, and the N-channel FET is on, so the output voltage falls to zero. Therefore, the circuit is a logic inverter: a low input results in a high output, and vice versa. In either logic state one FET is ON while the other is OFF. Because one FET is always turned OFF, the quiescent current of the the CMOS unit is extremely low. These properties of N- and P-type enhancement-mode FET's combined to form CMOS gates provide many advantages:  Extremely low power consumption.  Wide power supply voltage range.  High DC noise margin.  High input impedance.  Wide operating temperature range. The diagram in Fig. 11-a illustrates standard CMOS metal-gate technology (74C/4000), but there are many other CMOS technologies including the high-speed silicon-gate HC, HCT, and FACT families. Another digital logic technology called BiCMOS takes advantage of the lower power consumption and higher integration density of CMOS, and the higher speed adn superior drive capability of bipolar transistors. Power MOSFET's: Power MOSFET's exhibit the properties of small-signal MOSFET's such as high-input impedance and voltage control, and they have drains, sources and gater, but they are designed to handle higher currents. As majority-carrier devices that store no charges, they can switch faster than bipolar power transistors. Figure 12 is a section view of an N-Channel, enhancement-mode power MOSFET. Unlike its small-signal counterpart, the latest power MOSFET's are fabricated with vertical rather than planar structures. They are made with the double diffused (DMOS) process, and they have conductive silicon (polysilicon) gates. The gate of this device is isolated from the source by a layer of insulating silicon oxide. When a voltage is applied between the gate and source terminals, an electric field is set up within the MOSFET. This field alters the resistance between the drain and source terminals, and it permits conventional current to flow in the drain in response to the applied drain circuit voltage. There are also P-channel, enhancement-mode power

MOSFET's in which conventional current flows in the opposite direction of the N-channel device. Fig. 13 is the schematic symbol for a DMOS enhancementmode, N-channel power MOSFET. Figure 14 is an cutaway view of a typical DMOS power MOSFET. It is made up of many cells or transistor element connected in parallel. Each source cell consists of a closed rectangular or hexagonal channel which separates a source region from the substrate drain body. The cells are formed in an integrated circuit process, and there might be more than a half million cells per square inch of substrate. All of the source cells are connected in paralleled by a continuous deposition of aluminum metallization, which forms the grid-like common source terminal. The DMOS power MOSFET contains an inherent PN junction diode, and its equivalent circuit can be considered as a diode in parallel with the source-to-drain channel, as shown in the schematic symbol of Fig. 13. International Rectifier (IR) makes DMOS power MOSFET's that have hexagonally shaped cells, so it calls its products HEXFET's. Motorola Semiconductor also offers DMOS products MOSFET's, but its devices have rectangular rather than hexagonal cells. Motorola named its power MOSFET's TMOS to call attention to the T-shaped current flow that occurs in the cells between the common drain and the channels to the multiple sources. Power MOSFET's are widely specified for high-frequency switching power supplies (generally those that switch at frequencies above 100kHz), AC and DC motor speed controls, high-frequency generators for induction heating, ultrasonic generators, audio amplifiers, and amplitude modulation transmitters. The advantages to using power MOSFET's over power bipolar transistors include:  Faster switching speeds and lower switching losses.  Absence of the bipolar's second breakdown.  Wider safe operating area.  Higher input impedance.  High, if not higher, gain.  Faster rise and fall times.  Simple drive circuitry. The principal disadvantages of power MOSFET's are their higher cost and a higher static drain-to-source on-state resistance, which can cause unacceptable power losses in certain switching applications.. However, the manufacturers have made progress in reducing those resistance values. DMOS geometry had largely replaced the V-groove or VMOS

process that was widely used to fabricate power MOSFET's back in the 1970's. Radio-Frequency power MOSFET's are now available that will operate over the 2 to 200 MHz frequency range. The high power and high gain of these devices makes them suitable as power amplifiers in solid-state transmitters for FM and TV br5oadcasting. Copyright and Credits: © 1993 Original author Ray Marston. Electronics Now, March 1993. Published by Gernsback Publishing. (Gernsback Publishing is (sadly) out of business since January 2000). All graphics, drawings, photos, © 2006 Tony van Roon. Re-posting or taking graphics in any way or form from this website or of this project is expressly prohibited by international copyright laws. Permission by written consent only. Continue with Transistor Tutorial Part 10: "MOSFet's" Copyright © 2006 - Tony van Roon, VA3AVR Last updated: November 6, 2007